Ⅳ. Implementation of multi-physiological sensing ROIC
4.7 Experimental results of the ROIC
4.7.1 Experimental results of the biopotential AFE
Figure 4.32(a) shows an input-referred noise of the low-noise amplifier in the biopotential analog front-end. It is confirmed that flicker noise is dominant in a low frequency band, and thermal noise is dominant in a frequency above 40Hz. Integrated noise from 0.5Hz to 100Hz has a value of 0.69μVrms. Figure 4.32(b) shows a CMRR performance of the LNA, where a differential gain and common-mode gain were measured in the frequency range of 0.2Hz to 200Hz, and the difference between the two results was obtained by the CMRR. A mismatch condition of the electrode impedance was set by connecting a 1k resistor to an input (ZE1) and the 1k~100k resistor to the opposite input (ZE2). When the ZE2 is 1k, the CMRR performance of 99dB to 102dB is achieved, and it is confirmed that the CMRR decreases due to the electrode impedance mismatch.
In order to confirm the effectiveness of the proposed attenuator-assisted DSL, figure 4.33 shows the comparative measurements of the low-noise amplifier with the attenuator activated and deactivated.
Figure 4.33(a) shows frequency responses which are measured in the 0.2Hz to 200Hz frequency range and have a same passband gain of 41dB on the all conditions. With a 0.75kHz of duty-cycled resistor clock frequency (fDSL) on the DSL integrator, the HPF cutoff frequency is 1.4Hz when the attenuator is deactivated, but the HPF cutoff frequency is lower than 0.2Hz when the attenuator is activated. In order to identify how much the attenuator lowers the HPF cutoff frequency, the fDSL is increased to 15kHz,
Figure 4.32. (a) Input-referred noise and (b) CMRR performances of the biopotential AFE.
(a)
(b)
0.69μVrms [0.5-100Hz]
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and at this condition, the frequency response curve is almost same with the condition of attenuator deactivation. It means that the proposed attenuator-assisted DSL has the effect of lowering the HPF cutoff frequency by 20 times. On the other hand, it is necessary to have a high HPF cutoff frequency for the fast-settling mode in digital DSL operation. For this, the fDSL is increased to 75kHz and the HPF cutoff frequency is changed to about 10Hz.
Figure 4.33(b) shows the input and output waveforms with a normalized amplitude when the ECG and PPG waveforms which are repeated every second are applied as input signals. The signals are measured using two channels of the biopotential, and each input is generated using a function generator.
It is measured on the activated and deactivated conditions of the attenuator with 0.75kHz of fDSL. In the Figure 4.33. (a) Frequency response of the LNA and (b) transient waveforms under the activation and
deactivation of the proposed attenuator.
(a)
(b)
Without the attenuator, the peak shifts to forward.
Pass-band gain = 41dB
fhp≈
1.4Hzfhp
<
0.2Hzfhp
≈
10Hz72
both conditions, the ECG peaks are measured on the same location because the ECG peak has a high frequency. However, the PPG signal is distorted and the PPG peak is shifted forward when the attenuator is deactivated, and this phenomenon is improved when the attenuator is activated. Since the peak shifting cause a significant error in measuring the pulse transit time, the attenuator-assisted DSL helps to prevent the signal distortion and obtain an accurate peak position.
Figure 4.34(a) shows the maximum cancellation range of input DC offset (IDO) of hybrid DSL, where a black and red lines are the results of measuring while increasing the IDO from -360mV to +360mV and decreasing the IDO from +360mV to -360mV at each. When the IDO has a value between -340mV and +340mV, LNA output offset did not occur because the whole IDOs were removed by hybrid DSL. But, when the IDO increases above 340mV or decreases below -340mV, LNA output offset occurs, which means it is out of IDO tolerance of hybrid DSL. As explained in Chapter 4.2.1.3, the LNA operates to additionally remove the IDO through digital DSL when the analog DSL cannot sufficiently remove the IDO. Therefore, the two graphs have hysteresis behavior in the increasing and decreasing states of the IDO.
Figure 4.34(b) shows the transient ECG waveform with a lead-off detection operation and the Figure 4.34. Measured result of (a) maximum cancellation range of input DC offset and (b) transient
ECG waveform with a lead-off detection operation using the proposed hybrid DSL.
(a)
(b)
The RLEAD of 270kΩ is compensated only by the analog DSL.
Maximun IDO cancellation range
RLEAD = 0Ω RLEAD = 270kΩ RLEAD = 800kΩ RLEAD = 2MΩ
DD-DSL = 16
DD-DSL = 19 DD-DSL = 25
The large RLEAD values are compensated by the analog and digital DSL.
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measurement setup was configured as shown in the figure 4.11. A DC lead-off current was fixed to 100nA, and the resistance between the input ports was replaced with a passive resistor of RLEAD, and the input ECG signal is assigned from a function generator and AC-coupling capacitor. As the RLEAD
changes from 0 to 270kΩ, input DC offset voltage of 27mV is generated and it is removed only by analog DSL. However, when the RLEAD is changed from 270kΩ to 800kΩ, the IDO voltage of 80mV which exceeds the IDO tolerance of the analog DSL is generated, so the digital DSL operates together.
Likewise, when the RLEAD is increased to 2MΩ, the IDO voltage of 200mV is generated, so digital DSL output (DD-DSL) increases even more. Assuming that the increased RLEAD means the increased electrode impedance, it was verified that the DD-DSL increases when the electrode impedance increases as the electrode contact deteriorates.
Figure 4.35 shows the measured frequency responses of the PGA which is designed with the programmable filter and gain characteristics. As shown in the equation (4.21), the switching frequency (fCLK) of the switched-capacitor type resistor is related to a pole frequency of a low-pass filtering characteristic in PGA operation, where the frequency responses have about the same pass-band gain of 20dB because the gain control bits (DGAIN<1:0>) was equally set to 00. In the figure 4.35(a), the switching frequency is adjusted from 2.5kHz to 20kHz, and it was confirmed that the pole frequency gradually increases according to the switching frequency. The switching frequency can secure a wider pass-band frequency by using a higher switching frequency, and the switching frequency was appropriately adjusted in order to sufficiently support the frequency band of the bio-signals but to remove the noise of the high frequency band. Figure 4.35(b) shows a programmable-gain characteristic of the PGA, and the gain control bits adjusts the size of PGA input capacitor (CIN). The measure gains
Figure 4.35. Frequency responses of the PGA by adjusting (a) a switching frequency and (b) gain control bits.
(a) (b)
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are approximately changed from 20dB to 40dB, where the LPF pole frequencies in each graph are almost the same because the switching frequency was equally set to 10 kHz.