Fig.2.4 (a) Equivalent circuit of Hirota's λ/4 line with smaller dimensions, including artificial resonance circuits. b) The final equivalent λ/4 transmission line circuit. Fig.3.2 A single-stage bandpass filter based on the generalized filter model (a) and its miniaturized form (b). Fig.3.4 Simulation results of 1- and 2-stage filters for skirt characteristics comparison: (a) Passband; (b) loss of efficiency.
그림 3.5 결합된 라인의 서로 다른 전기적 길이에 따른 서로 다른 대역폭: (a) 통과대역; (b) 반사 손실. 본 논문에서는 세로로 단락된 평행 결합선과 집중형 커패시터를 사용하여 새로운 소형 GaAs 대역통과 필터를 제안하였다. 그리고 설계된 GaAs 필터는 넓고 높은 저지대역을 갖습니다.
Introduction
In this paper, GaAs process based MMIC filter will be introduced for the RF single transceiver chip. The electrical length of resonator in MMIC filter can be reduced as small as a few degrees. Most chip filter using this concept can be designed to be less than 2´1 mm2.
This technology is available for any kind of standard manufacturing process because the topology of this filter circuit is only planar two-dimensional structures. Finally, it is also widely applicable from IF to millimeter band because the electrical length of it can be arbitrarily controlled. A filter using the GaAs process technology for a single transceiver chip is designed and manufactured at 5.5GHz to maximize the effect of the size reduction method, because the SAW filter covers the frequencies below 3GHz and the ceramic filter is still too large to insert in RF.
Size Reduction Method for the Quarter-wavelength
Introduction
The low-pass filter makes it possible to transmit low-frequency signals from the input to the output port with little attenuation. However, as the frequency exceeds a certain cut-off point, the attenuation increases significantly with the result of providing an amplitude-reduced signal to the output port. The opposite behavior applies to a high-pass filter, where the low-frequency signal components are strongly attenuated or reduced in amplitude, while the signal beyond a cut-off frequency point passes the filter with little attenuation.
Bandpass and bandstop filters limit the passband between specific lower and upper frequency points where the attenuation is either low (bandpass) or high (bandstop) compared to the remaining frequency band. Lump element filters are frequently applied in a low-frequency band because the wavelength of the operating signal will be comparable to the size of the lump elements themselves, while the distributed element filters can be used in a high-frequency band extending to tens and hundreds of gigahertz. Due to the variety and diversity of the filter types, it often becomes necessary for a designer to carefully consider which filter to adopt for a particular application.
Size Reduction Method
Hirota’s size reduction method for λ/4 transmission line
Equation (2.3) shows that the characteristic impedance Z increases as the electrical length θ decreases. So far, the limitation of the electrical length of the transmission line is about λ/8 ~λ/12. We can see from equation (2.6) that the shorted coupled lines are suitable for extremely miniaturized λ/4 transmission lines, since the high characteristic impedance can be easily achieved by choosing Zoe ≈ Zoo.
The high impedance transmission line with shunt lumped inductors can be replaced by coupled lines shown in Fig.2.4 (b). The peculiarity of this extremely miniaturized λ/4 transmission line is that resonant circuits are located at the edge side of the transmission line. When the miniaturized λ/4 transmission lines are connected in series, the cascade circuit becomes a typical bandpass filter with the λ/4 section as an access inverter.
The bandwidth of the filter can be controlled by the coupling coefficient since the bandwidth of diagonally shorted coupled line is closely related to the coupling coefficient [10].
Bandpass Filter Design Theory
Introduction
Although the characteristics of the inverter are relatively narrowband in nature, this quarter-wavelength line can easily be used as an access inverter in our proposed narrowband filters.
Size-reduced Bandpass Filter
Figure 3.3 shows the ADS model of the proposed one- and two-stage reduced-size filters. The 1- and 2-stage simulation results of the proposed bandpass filters are given in Figure 3.4 and here we have chosen 7 degrees as the electrical length of the coupled lines. The electrical length and the coupling coefficient of the terminally shorted lines are mainly two factors that can affect the bandwidth of the proposed bandpass filter.
In Fig.3.5, the quarter-wavelength transmission line is miniaturized to 3, 5, and 7 degrees, and the ADS simulation results fully proved the correctness of this discussion. From equation (3.2), the minus value in the parentheses indicates the 180° phase difference between the original quarter-wavelength transmission line and its miniaturized equivalent circuit, which was introduced when we equated the PI network with the shorted-pair lines at the end. At this point, the most important point is to reduce the frequency sensitivity of the equivalent transmission line l/4.
If Y0e is small, the frequency sensitivity decreases due to the very small shunt value of C0w in the artificial resonant circuit. The larger the coupling coefficient K, the wider the bandwidth of the bandpass filter, as shown in fig. 3.6, where the electrical length of the coupled lines is chosen to be 7 º and two steps are used.
Before concluding the design theory of bandpass filter, here a comparison is made between the proposed bandpass filter and the traditional one. At the same time, the insertion of the short transmission line does not affect the active characteristics of the modified bandpass filter and has a good effect on the result. A very small length of the input and output transmission line will lead to a very small inductance and therefore an extremely large impedance, and the input signal will not be able to flow at all.
At the same time, two more poles are added due to the added resonant circuits, which contribute to the sharp shirt characteristics of the modified bandpass filter. For a given Z0, the electrical length of the paralleled lines can be made very small as long as we choose Z0e » Z0o. As shown above, the main key factors affecting the bandwidth of the bandpass filter are the electrical length and the coupling coefficient of the short-terminated coupled transmission line.
Simulation and Measurement Results
In the extremely miniaturized circumstances, it is very difficult to manufacture each component exactly as designed due to unexpected coupling between components. Subsequently, we simulated the circuit with a soft HFSS V9 to obtain the effect of the overall response. According to Fig.4.6, the effects of eight through holes in the filter circuit can be ignored.
In Figure 4.3, there is an 80 um interstage transmission line to prevent unexpected coupling between two resonators. Although HFSS was used to find the true value of the capacitor in the simulation part, the MIM capacitor was used in the HFSS model for simulation. The capacitance of the lumped capacitor available in the laboratory is slightly different from that of the MIM capacitor, which leads to a slightly deviated center frequency.
This can be expected due to the boundary and excitation setting in HFSS, the inserted transmission line between the phases and the. But if we cascade it to a two-stage circuit, it will probably be a little different than the single-stage circuit because of the size of the transmission line between the stages. The number of attenuation poles is different after changing the size of the interstage transmission line.
Compared with the measurement result, there is a little difference between them due to design and simulation errors, the accuracy of the manufacturing process, and artificial errors in the measurement. Fig.4.6 compares the measured data with the HFSS simulated results, and it is clear that there is good agreement with the two results. In the measurement results, the passband has a maximum insertion loss of 6.5dB with 0.9GHz bandwidth, from 4.8GHz to 5.71GHz and 13dB return loss.
This ultra-wide stopband characteristic is a particular advantage compared to ceramic or SAW filters.
Conclusion
Denidni, “A new compact microstrip two-layer bandpass filter using aperture-coupled SIR hairpin resonators with transmission zeros,” IEEE Trans. Kakimoto, “Miniaturized hairpin resonator filters and their application to receiver front-end MICs,” IEEE Trans. Lancaster, “Theory and experiment of new open-loop microstrip slow resonator filters,” IEEE Trans.
Chang, “Realization of transmission nulls in combinational filters using an inductively coupled auxiliary ground plane,” IEEE Trans. I would like to express my heartfelt gratitude to the many people who have helped me both in my studies and in my daily life during the past two years in Korea. Without their generous support and help, it would have been impossible for me to finish the study, even the dissertation.
I cannot be grateful enough to my advisor, Professor In-ho Kang, whose wide knowledge, rigorous research attitude and enthusiasm in work deeply impressed me and taught me what a true scientific research should be. I am also grateful to the other professors of our department for their support and guidance on this work, which are Professor Dong Il Kim, Professor Kyeong-Sik Min, Professor Hyung Rae Cho, Professor Ki Man Kim, Professor Ji Won Jung, Professor Young Yun, Professor Joon hwan Shim and Professor Dong Kook Park in the Department of Electronics and Communication Engineering. They not only help me with my study work, but also let me enjoy the friendly working environment.
My deepest thanks also go to my friends in the Microwave and Antenna Laboratory, Mr.
