MTI FROM A MOVING PLATFORM

MTI AND PULSE DOPPLER RADAR

4.11 MTI FROM A MOVING PLATFORM

When the radar itself is in motion, as when mounted on a ship or an aircraft, the detection of a moving target in the presence of clutter is more difficult than if the radar were stationary. The doppler frequency shift of the clutter is no longer at de. It varies with the speed of the radar platform, the direction of the antenna in azimuth, and the cl.evation angle to the clutter. Thus the clutter rejection notch needed to cancel clutter cannot be fixed, but must vary. The design of an MTI is more difficult with an airborne radar than a shipborne radar because the higher speeds and the greater range of elevation angles result in a greater variation of the clutter spectrum.

In addition to shifting the center frequency of the clutter, its spectrum is also widened. An approximate measure of the spectrum width can be found by taking the differential of the doppler frequency .f.i =

2(v/l)

cos 0, or

4fd = 2

; sin O !lO (4.36)

where v = platform speed, l = wavelength, and O is the azimuth angle between the aircraft's velocity and the direction of the antenna beam. (The negative sign introduced by differentia- tion of cos O is ignored and the elevation angle is assumed to be zero.) If the beamwidth is

taken as MJ, then

M~

is a measure of the width of doppler frequency spectrum. When the antenna points in tile direction of tile platform velocity (0 = 0), the doppler shift of the clutter is maximum, but the width of the doppler spectrum llfd is a minimum. On the other hand, when the antenna is directed perpendicular to the direction of the platform velocity (0 = 90°), the clutter doppler center-frequency is zero, but the spread is maximum. This widening of the clutter spectrum can set a limit on the improvement factor.

Thus the elTect of platform velocity can be considered as having two components. One is in the direction of antenna pointing and shifts the center frequency of the clutter doppler spectrum. The other is normal to the direction of antenna pointing and results in a widening of the clutter doppler spectrum. These two components are compensated by two different techniques.

J\n MTI radar on a moving platform is called AMTI. Although the "A" originally stood for airborne, the term is now often applied to an MTI radar on any moving platform. Most of the interest in AMTI, however, is for airborne radar.⁵⁸

Compensation for clutter doppler shift. When the clutter doppler frequency is other than at de, the null of the frequency response of the MTI processor must be shifted accordingly. The effect on the improvement factor when the center of the clutter doppler frequency is shifted by an amount./~ is shown in Fig. 4.34 for a three-pulse delay-line canceler.⁶¹•69 There are two basic methods for providing the doppler frequency compensation. In one implementation the frequency of the coherent oscillator (coho) is changed to compensate for the shirt in the clutter doppler frequency. This may be accomplished by mixing the output of the coho with a signal from a tunable oscillator, the frequency of which is made equal to the clutter doppler. The other implementation is to insert a phase s~ifter in one branch of the delay-line canceler and adjust its phase to shift the null of the frequency response. (A phase shift \JI in one branch of the canceler corresponds to a frequency shift 2nfd

=

\JI /Tp, where TP = pulse repetition interval.) The clutter-doppler-frequency compensation can, in some cases, be made open loop by using the a priori knowledge of the velocity of the platform carrying the radar and the direction of the antenna pointing. This is more practical with a shipborne radar rather than with an airborne radar. When the clutter-doppler-frequency compensation cannot be obtained

100

90 m "O

!? ...

u

2

c

_Q)

E Q)

>

0

a

E 10

0

0001 001 0·1

Uclfp

Figure 4.34 Effect of a nonzero clutter doppler frequency on the improvement factor of a three-pulse canceler. fr= mean frequency of the clutter spectrum, u, = standard deviation of clutter spectrum,

fP

=

pulse repetition frequency. (From Andrews.⁶¹)

in this fashion, the clutter frequency can be measured directly by sampling the received echo signal over an interval of range. The sampled range interval is selected so that clutter is likely to be the dominant signal. From this measurement of clutter doppler within the sampled range interval, compensation is made over the entire range of observation either by changing the reference signal from the coho or by adjusting the phase shifter inserted in one of the arms of the delay-line canceler. Generally, the average doppler frequency or phase shift is obtained by averaging the sampled range-interval over a number of pulse repetition periods. A single doppler measurement and subsequent compensation might not suffice over the entire range of the radar, especially if the radar is elevated as in an aircraft. The doppler shift from clutter will be range dependent with an elevated radar since the doppler frequency is a function of the elevation angle from the radar to the clutter cell. Thus more than one doppler measurement may be necessary to compensate for the variation of the clutter doppler with range.

An MTI radar that measures the average doppler frequency shift of clutter qver a sampled range interval and uses this measurement to cause the clutter mean-doppler-frequency to coincide with the null of the MTI doppler-filter-frequency response over the remainder of the range of observation is called a clutter-lock MT/.

One version of a clutter-lock MTI is TACCAR, which stands for time-averaged-clutter coherent airborne radar.⁸ Although the name was originally applied to a particular airborne MTI radar system developed by MIT Lincoln Laboratory, it continues to be used to refer to the clutter-lock technique that was the special feature of that system. The name is even retained when the technique is applied to a shipboard radar or when the radar is on land and used for the compensation of moving clutter. The chief feature of TACCAR is the use of a voltage-controlled osci11ator arranged in a phase-lock loop. As with other clutter-lock methods, the correction for the clutter doppler is obtained from the averaged measurement of the clutter doppler frequency within a sampled range interval.

Other methods that have been proposeci for compensating the clutter doppler shift seen by a moving radar include the "matrix MTI "⁶⁰ for implementation in digital MTI, and a

"trial and error" tech~ique^59•68 that provides simultaneously a number of possible doppler corrections and uses that

one

which produces the minimum residue over the sampled range interval.

Still another technique that is attractive when it can be applied, is to use a doppler processor with a rejection notch wide enough to reject the clutter doppler even when the radar platform is in motion. This is applicable only when the first blind speed is high (a low radar frequency and/or high prf) and when the platform speed is low, as it would be with a ship- mounted radar.

Generally, most clutter-lock MTI techniques· do not adequately eliminate both stationary and moving clutter when they appear simultaneously within the same range resolution cell. A TACCAR, for example, might be designed to reject ground clutter close in, and weather or chaff at a different doppler at ranges beyond the ground clutter; but not to cancel two different clutter doppler frequencies simultaneously.⁸ An exception is the adaptive MTI^62•65 which can adapt to any type of clutter. Using technology similar to that of the antenna sidelobe canceler, nulls are adaptively placed at those frequencies containing large clutter. A three-loop adaptive canceler, for example, can adaptively place three nulls at three different frequencies or it can place the three nulls so as to make a single wide notch, depending on the nature of the clutter spectrum.

Compensation for clutter doppler spread. The simple expression of Eq. (4.36) shows that the spread in the clutter spectrum is a function of the angle 8 betwC?Cn the velocity vector of the moving platform and the antenna beam-pointing direction. It also depends on the wavelength.

Based on an analysis of antenna radiation patterns and experimental data, Staudaher⁵⁸ gives the standard deviation of the clutter spectrum due to platform motion as

'1pm ::::::0: 0.6 Vx

a

^(4.37)

where vx is the horizontal component of the velocity perpendicular to the antenna pointing- direction and a is the effective horizontal aperture width. The antenna beamwidth is assumed to be approximated by 08

=

},_/a. [Equation ( 4.3 7) is not inconsistent with the simpJe derivation of Eq. (4.36 ).] If the mean doppler-frequency shift of the clutter echo is perfectly com- pensated, the limitations on the improvement factor due to clutter spread can be found by assuming a gaussian spectral shape and substituting the standard deviation of Eq. (4.37) into the expression of Eq. (4.27) to obtain

I pm = ( ; (

l.~n ~~~jJ

^2N1

^(4.38)

where N₁ is the number of delay lines in the MTI processor. If the clutter spread cr₁₁mdue to the platform motion combines with the clutter spread <le due to internal clutter motion such that the total standard deviation <rr of the clutter spectrum is o-} = a-;

+ o-:m,

the MTI improve- ment factor for the total clutter spectrum is

(4.39) The solid curves of Fig. 4.35 plot this equation for a three-pulse delay~line canceller (N₁

=

^2).

If the widening oft he spectrum is a result of the radar platform·s velocity, its effects can be mitigated by making the radar antenna appear stationary. This might be accomplished with

(D

u

...

0

u 0

...

c

_ID

E «>

>

0

f 0.

80 - - -

-- -- _--

^...

^_

X"" 0 01 - -

X::: Q 01

---

60

----

^~

40

20

X

=

01

o~~~~~~~~~~~~~~~---'-~~~~~~~~~~~~~~~~

0001 0·01 0·1

Ratio of clutter spectral width lo pulse repetition frequency (o-c lfp)

Figure 4.35 Solid curves show the improvement factor of a three-pulse canceler limited by platform motion [Eq. (4.39)]. Dashed curves show the effect

or

the DPCA compensation. x is the fraction of the antenna aperture that the antenna is displaced per interpulse period (x = 0 corresponds to no platform motion.) (From Andrew.s.⁶³⁾

two separate antennas with the distance between them equal to Tpl'x

=

TPv sin Oa, where Oa

=

angle between velocity vector of the vehicle and the antenna beam-pointing direction, and TP

=

pulse repetition period. Orie pulse is transmitted on the forward antenna, and the other pulse is transmitted on the rear antenna so that the two pulses from the two different antennas are transmitted and received at the same point in space. The result is as if the radar antenna were stationary. The distance traveled between pulses is generally less than the antenna dimension so that the two antenna beams might be generated with two overlapping reflector antennas or with a phased array divided into two overlapping subarrays.⁶⁴ The effective separ~tion between the antennas, T pvx, varies with the angle 0₀ as well as the aircraft velocity v. With reflector antennas, it is not convenient to change the antenna physical separation to compensate for changes in Oa or v. The pulse repetition period TP might be varied to provide compensation, but this can introduce other complica- tions into the radar design and the signal processing. With a phased array divided into two overlapping subarrays, a constant pulse repetition frequency can be used and the horizontal separation of the two overlapping subarrays can be controlled electronically to compensate for platform motion. However, it is possible to change the effective phase center of a rcllcctor antenna by employing two feeds to produce two squinted overlapping beams, as in an amplitude-comparison monopulse radar (Sec. 5.4). The outputs of the two feeds are combined using a hybrid junction to produce a sum pattern I: and a difference pattern L\.

By

taking I:

±

jkL\, the effective phase center can be shifted depending on the value of k. (The factor j multiplying the difference pattern signifies a 90° phase shift added to the difference signal relative to the sum signal.) The use of this technique in an AMTI radar to compensate for the effects of platform motion is called DPCA₁ which stands for Displaced Phase Center Antenna.

The sum and difference patterns can be obtained by connecting a hybrid junction to the outputs of the two antenna feeds as described in Sec. 5.4. The sum pattern is used 0:1 transmit and both the sum and the difference patterns are extracted on receive. The signal received on the difference pattern is weighted by the factor k, shifted in phase by 90° and is added to the sum-pattern signal in the delayed channel and subtracted from the sum-pattern signal in the undelayed channel. Because of the phase relationships between the lobes of the difference pattern and the sum pattern, the resu1t is an apparent forward displacement of the pattern on the first transmission, and a displacement to the rear on the second transmission. When th~

gains of the sum and difference channels are properly adjusted, and when the distance between the phase centers of the two antenna beams is 2Tpvx the combined sum and difference p~tterns on successive pulses illuminate the same region and the antenna appears stationary. ⁵¹¹ (The factor 2 appea·rs in the distance between phase centers. as a result of using both feeds for transmission. The phase center on transmit is half-way between the two feeds, and the phast:

center on receive alternates from one feed to the other.) As the antenna pointing-direction changes from the port to starboard side of the vehicle, the sign of the difference signal must be reversed to keep the displaced beams in the proper orientation.

The dashed curves of Fig. 4.35 show the improvement in MTJ processing that is theo- retically possible with DPCA and a three-pulse delay-line canceler. (Note that the DPCA corrects only one canceler of a multiple-stage MTl.⁶³) The curve for x = 0 applies for no platform motion and represents the maximum improvement offered by an idea platform- motion cancellation method. It is seen that when the clutter spectral width is small, as for overland clutter, a significant improvement is offered by DPCA.

The limitation to the improvement factor due to antenna rotation, or scanning modula- tion, can be redu~ed

by

a method.similar to DPCA.⁵¹•58

•66 DPCA applies the difference pattern in quadrature (90° phase shift) to the sum pattern while compensation for scanning modulation requires the difference pattern to be applied in phase with the sum pattern. Thus

it

is possible to combine the two techniques for compensating platform motion and scanning modulation.⁵⁸

If the antenna sidelobes of an airborne MTI radar are not sufficiently low, the clutter that enters the receiver via the sidelobes can set a limit to the improvement factor equal to⁵⁸

_ K

f " ^G

²

⁽⁰⁾

^dO

/~,- -J-G²(0)d0

sl

(4.40)

where G(O) is the one-way power gain of the antenna in the plane of the ground surface. The lower integral is taken outside the main-beam region. This assumes the sidelobcs are well distributed in azimuth. The constant K is the average gain of the delay line canceler (K

=

2 for a two-pulse canceler and 6 for a three-pulse canceler.) The combined improvement factor for DPCA and the sibelobe limitation is

+

l

I~, l nrcA

(4.41)

Adaptive array antennas may be employed to compensate for platform motion in an AMTI radar.^6' The full array is illuminated on transmission so that the transmit pattern is the same from pulse to pulse. On receive, the array is made adaptive by obtaining a separate output from each element. Each element output can be weighted separately and the outputs added together to form an aperture illumination function that adaptively permits motion

or

the antenna phase center so as to compensate for platform motion. Adaptive loops are also used in the delay-line canceler to control the doppler response of the canceler as well as the antenna angular response. In addition, compensation for scattering from near-field aircraft structure that distorts the antenna pattern and degrades AMTI performance can also be performed with this adaptive circuitry, as can the adaptive nulling of external interference sources.

For applications that cannot afford a fully adaptive array antenna, a design procedure can he formulated that applies an optimal correction to an arbitrary receive array antenna pattern hased on the use of a least-mean-squares algorithm to minimize the total clutter residue of an I\ MTI radar averaged over all angtes.⁷⁴

Sidelobes and pulse-doppler radar. Since the pulse-doppler radar is capable of good MTI performance, it is also a good AMTI radar. However, if the antenna sidelobes are not low, the clutter that enths the radar via the sidelobes can limit the improvement factor, as mentioned previously [Eq. (4.40)]. The effect of the sidelobe clutter must often be considered in the design of the signal processor of an airborne pulse-doppler radar.

The spectrum of the signal received by an airborne pulse-doppler radar might appear as in Fig. 4.36. Only that portion of the spectrum in the vicinity of the carrier frequency

lo

is shown since the prf of a pulse-doppler radar is chosen to avoid overlap of target signals from adjacent spectral lines (no blind speeds). Thus the prf is at least twice the maximum target doppler-frequency. The leakage of the transmitter signal into the receiver produces the spike at a frequency

lo

and the spikes at Jo

± ,,JP

where

n

is an integer and

JP

is the pulse repetition frequency. Also in the vicinity of

lo

is the clutter energy from the sidelobes which illuminate the ground directly beneath the aircraft. The echo from the ground directly beneath the aircraft is called the altiwde return. The altitude return is not shifted in frequency since the relative velocity between radar and ground is essentially zero. Clutter to either side of the perpendicu- lar will have a relative-velocity component and hence some doppler frequency shift; con- sequently the clutter spectrum from the altitude return will be of finite width. The shape of the

~ Transmitter to receiver leakage Altitude return-...,..

Frequency

,,,, Main-lobe clutter

y

,.-Target echo (head-on)

fo+fc

t

^fo+fd

f₀+2v/A

Figure 4.36 Portion of the received signal spectrum in the vicinity of the RF carrier frequency

h.

for a pub doppler AMTI radar. ( After M aquire, ⁷⁰ Proc. Natl. Conj on Aero11aut. Electro11ics.)

altitude-return spectrum will depend upon the variation of the clutter cross section as a function of antenna depression angle. The cross section of the clutter directly beneath the aircraft for a depression angle of 90° can be quite large compared with that at small depression angles. The large cross section and the close range can result in considerable altitude return. (_)₁

The clutter illuminated by the antenna sidelobes in directions other than directly beneath the aircraft may have any relative velocity from

+

v to - v, depending on the angle made by the antenna beam and the aircraft vector velocity (v is the aircraft velocity). The clutter spectrum contributed by these sidelobes will extend 2v/). Hz on either side of the transmitter frequency. The shape of the spectrum will depend upon the nature of the clutter illuminated and the shape of the antenna sidelobes. For purposes of illustration it is shown in Fig. 4.36 as a uniform spectrum.

The altitude return may be eliminated by turning the receiver off (gating) at that range corresponding to the altitude of the aircraft. Gating the altitude return has the disadvantage that targets at ranges corresponding to the aircraft altitude will also be eliminated from the receiver. Another method of suppressing the altitude return in the pulse radar is to eliminate the signal in the frequency domain, rather than in the time domain, by inserting a rejection filter at the frequency .fo. ,The same rejection filter will also suppress the transmitter- to-receiver leakage. The clutter energy from the main beam may also be suppressed by a rejection filter, but since the doppler frequency of this clutter component is not fixed, the rejection filter must be tunable and servo-controlled to track the main-beam clutter as it changes because of scanning or because of changes in aircraft velocity.

The position of the target echo in the frequency spectrum depends upon its velocity relative to that of the radar aircraft. H the target aircraft approaches the radar aircraft head on (from the forward sector), the doppler frequency shift of the target will be greater than the doppler shifts of the clutter echoes, as shown in Fig. 4.36. A filter can be used to exclude the clutter but pass the target echo. Similarly, if the targets are receding from one a not her along headings 180° apart, the target doppler frequency shift will again lie outside the clutter spec- trum and may be readily separated. from the clutter energy by filters. In other situations where the radar may be closing on the target from the tail or from the side, the relative velocities may be small and the target doppler will.lie within the clutter doppler spectrum. In such situations the target echo must compete with thedutter energy for recognition. A large part of the clutter energy may be removed with a bank of fixed narrowband filters covering the expected range of doppler frequencies. The bandwidthof each individual filter must be wide enough to accept the energy contained in the target echo signal. The width of the filter will depend upon the time on target, equipment fluctuations~ and other effects which broaden the echo-signal spectrum as discussed previously_. Each of the doppler filters can have its own individually set threshold whose level is determined by the amount of noise or clutter within the filter. This can be done