Chapter 4 Concurrent Multi-Band Amplifiers 40
5.1 Top-Level Radio Design
5.1.2 Receiver Architecture
PN(foffset)[dBc/Hz] ≤ C[dBm] – 10log[C/(I+N)] – I[dBm] – 10log(B) (5.2) In (5.2), C is the signal level that has to be recovered in the presence of interference level I at frequency foffset.
The target receiver dynamic-range, NF, and VCO phase-noise of the standards in Table 5.1 are derived in Table 5.2. For comparison purposes, the numbers for the more stringent GSM standard for mobile-phone communication are also mentioned in Table 5.2.
Table 5.2: Receiver target numbers for standards in Table 5.1
Once again, it should be noted that the implementation of the concurrent receiver is most valuable in demonstrating the new concept than in satisfying the application-specific radio requirements. Nonetheless, Table 5.2 provides realistic performance goals in designing the radio.
Figure 5.1: Complete schematic of the fully integrated concurrent dual-band receiver
The received signals at two frequency bands are first amplified by a concurrent dual- band low-noise amplifier. The narrow-band response of the LNA around each of the desired frequency bands prohibits the amplification of strong out-of-band signals that might deteriorate the performance of later stages in the receiver chain due to nonlinearity. In a receiver, the gain of each block suppresses the effect of noise of the following stages in the overall performance [46]
⋅⋅
⋅
− +
− + +
=
2 1 3 1 1 2
1 1
G G F G F F
Foverall (5.3)
In (5.3), Foverall is the overall noise-factor of the receiver, and Fi and Gi are noise-factor and available gain of the ith stage in the receiver chain.
However, as the signal is amplified along the receiver, its amplitude gets larger and is more likely to drive the following stages into undesired nonlinear region of operation, i.e.,
⋅⋅
⋅ + + +
=
3 2 1 2 1
1 3 3
3 1 3
1
IIP G G IIP
G IIP
IIP overall (5.4)
Concurrent Dual-Band
LNA
Q
Low Band (frf=2.45GHz)
90o
Loop Filter Charge
Pump
Concurrent IF Amplifier
PFD
÷2
÷256
fref≈12.3MHz
VCNTRL
÷3
÷3
90o 90o
I Q
High Band (frf=5.25GHz)
I Q
6.3GHz 2.Cd
M3
0.7GHz flow-band = 2.45GHz
fhigh-band = 5.25GHz
Q
In (5.4), IIP3overall is the input-referred intercept point of the complete receiver, and IIP3i
and Gi and the input-referred intercept point and linear gain of the ith stage in the receiver chain.
Therefore, the gain of LNA should be large enough to suppress the noise of later stages and low enough not to sacrifice the linear behavior of following stages. In order to relax the dynamic-range requirements of the receiver, two gain modes for the concurrent dual-band LNA are devised. When the received signal strength is low (close to sensitivity level) the amplifier has a larger gain. In the case of a large received signal, the amplifier gain is reduced to provide a reasonable level to next stages. In the concurrent case, it is desired to have an independent control of gain for all the desired frequency bands. However, in this implementation of concurrent LNA a single gain-control is applied for both frequency bands.
Assuming symmetric in-phase and quadrature-phases signal paths, the down- conversion scheme proposed by Weaver [8] rejects the image signal completely.
Nevertheless component mismatches inherent to any actual implementation, lowers the image-rejection of Weaver architecture considerably. In fact, it can be shown that a mismatch of ∆G and of ∆Φ in the gain and phase of in-phase and quadrature paths sets the following limit on the image-rejection (IMR) of the receiver [35],[36].
( )
4 2
2
∆ +
∆Φ
= G
G
IMR (5.5)
In order to reach a higher on-chip image-rejection, quadrature mixing in the radio frequency (RF) mixers followed by the use of double quadrature downconversion scheme ([100]) at the intermediate frequency (IF) stage is pursued. The use of complex mixing enhances the image-rejection by an order of magnitude [100]. Furthermore, an even larger on-chip image-rejection is achieved by a combination of frequency planning and the best use of the notch in the dual-band front-end transfer function, as mentioned in Section 3.2.6.
The complete frequency planning of the receiver with the frequency details is illustrated in Figure 5.2. As evident from this figure, since the received RF signals are down-converted into two different intermediate frequencies, a dual-band response at the IF stage is considered as well.
Figure 5.2: Frequency planning of the concurrent dual-band receiver in Figure 5.1
Since high-frequency PLLs consume a considerable amount of battery power and chip area, it was decided to generate all the local oscillator frequencies out of one PLL and high-speed dividers. Channel selection can be achieved at baseband with substantially smaller power consumption.
Compared to other dividers, divide-by-two circuits are generally considered to be easier to design. Additionally, both in-phase and quadrature-phase signals can be extracted from the outputs of divide-by-two and divide-by-four circuits [102]. Hence, a frequency planning scheme in which all the local-oscillator signals are generated from a single oscillator and divide-by-two blocks was conceived (Table 5.3).
Table 5.3: Concurrent receiver frequency planning that uses only divide-by-two
fLO1 fLO2=fLO1/2 fLO3=fLO1/4
3.42GHz 1.71GHz 0.855GHz
fRF1 fIF1=fLO1-fRF1 fimg1=fIF1+fLO1 fBB1=fIF1-fLO3
2.45GHz 0.97GHz 4.39GHz 115MHz
fRF2 fIF2=fRF2-fLO1 fimg2=fLO1-fIF2 fBB2=fIF2-fLO3
5.25GHz 1.83GHz 1.59GHz 120MHz
f
0 f
A
0 f
B
Desired Bands Image Bands Dual-Band Front-End Transfer Function
3.85
2.45 3.15 5.25 GHz
1.05
f
0.70 2.10 GHz
Dual-band Mixer/IF 6.3-GHz
PLL
÷2 ÷3
3.15-GHz ÷3
0.7-GHz 2.1-GHz
The downfall of the aforementioned frequency planning is the closeness of image frequency of each band (e.g., fimg1) to the center frequency of the other band (e.g., fRF2).
Consequentially, dual-band transfer function of the front-end can not attenuate the image- signals as much20.
Improved frequency planning schemes can be considered that utilize frequency divisions by an integer factor other than two21. A frequency planning scheme that exploits a divide-by-three and two divide-by-two blocks to generate all the necessary local-oscillator signals from a 6.6 GHz source is presented in Table 5.4.
Table 5.4: Concurrent receiver frequency planning that exploits divide-by-three and divide- by-two circuit blocks
Compared to the previous case, the image frequency of each band is farther away from the center frequency of the other band that allows for a better image-rejection due to the dual- band front-end transfer function. The final down-converted signals are centered at 250 MHz that will necessitate a wide-band digital circuitry for final channel selection and demodulation.
The frequency plan of Figure 5.2 was selected for the implemented receiver for a variety of reasons. A single PLL at 6.3 GHz and its divided versions can generate all the frequencies required in the receiver. Since the first local oscillator is generated by a divide- by-two, we will have access to both in-phase and quadrature phases of the LO. A passive polyphase filter ([100]) further corrects for any phase imbalance between the two that could
20 It should be mentioned that placing the front-end notch very close to either of the frequency bands results in a lower gain and bandwidth for that band.
21 In many cases, analog frequency multiplication can be exploited instead of frequency division.
fVCO fLO1= fVCO/2 fLO2=fLO1/3 fLO3=fLO2/2
6.6 GHz 3.3 GHz 2.2 GHz 1.1 GHz
fRF1 fIF1=fLO1-fRF1 fimg1=fIF1+fLO1 fBB1=fIF1-fLO3
2.45 GHz 0.85 GHz 4.15 GHz 250 MHz fRF2 fIF2=fRF2-fLO1 fimg2=fLO1-fIF2 fBB2=fIF2-fLO3
5.25 GHz 1.95 GHz 1.35 GHz 250 MHz
be due to the potential asymmetric duty-cycle of the 6.3 GHz oscillator. The second and third local oscillator signals are generated with divide-by-threes and are passed through passive polyphase filters for quadrature generation. Generation of in-phase and quadrature- phase signals using polyphase filters will be further discussed in subsection 5.2.4. The image frequencies are located far enough from the frequency bands of interest and will be attenuated by the front-end transfer function substantially. Additionally, the desired signals are down-converted to dc or a very low-IF and can be further processed in digital domain.
The strength of the received signal can be measured at baseband and a feedback circuitry can change the gain of low-noise amplifier and other low-frequency variable-gain amplifiers that are not a part of described implementation. In summary, a large dynamic- range at both frequency bands should be expected from the concurrent dual-band receiver in Figure 5.1.