Bidirectional Converter in Discontinuous Conduction Mode with Zero Voltage Switching Control and Variable Switching Frequency
1Someswari T, 2Pooja H K
1,2Department of EEE, TOCE, Bangalore.
Abstract— This project investigates a control approach for achieving reliable zero-voltage switching transitions within the entire operating range of a conventional no isolated bidirectional DC-DC converter that utilizes synchronous rectification. The approach is based on operation in the discontinuous conduction mode with a constant reversed current of sufficient amplitude, which is achieved by load-dependent variation of the switching frequency. This project focuses on the obtained resonant voltage transitions and provides analytical models for determining the reversed current and timing parameters that would ensure safe, reliable and highly efficient operation of the converter. In addition, the proposed approach solves the synchronous transistor’s spurious turn-on and body diode reverse recovery induced issues, does not require any additional components or circuitry for its realization, and can be entirely implemented within a digital signal controller.
MATLAB simulation is carried out for the performance analysis and the hardware prototype is designed with microcontroller for pulse generation.
Index Terms— Bidirectional dc-dc converter, synchronous rectification, discontinuous conduction mode (DCM), zero- voltage switching (ZVS), resonant switching transitions, reversed current, reverse recovery induced issues
I. INTRODUCTION
One of the main advantages of synchronous dc-dc converters over conventional single-switch buck or boost converters is in the lower conduction losses achieved by replacing the freewheeling diode with a power transistor . The latter has a considerably lower internal resistance in the conductive state than a fast schottky diode and therefore lower conduction losses.
Another favorable property of the half bridge synchronous rectification structure is the possibility of a bidirectional power flow . In regard to this property such converters are widely used in different kinds of hybrid power systems and hybrid electric vehicles as chargers for the auxiliary energy storage units. Nonetheless, synchronous rectification also introduces several unwanted phenomena. A major concern in synchronous dc-dc converters is the reverse recovery effect of the synchronous transistor’s antiparallel body diode at the instant of turning on the main transistor (main FET). For the duration of the reverse recovery phenomena, both transistors are exposed to high current change rates and
high peak reverse recovery currents which significantly increase switching losses, cause electromagnetic interference, and lead to dangerous operating conditions.
Another drawback of hard-switched converters that utilize synchronous rectification is the unwanted Cdv=dt-induced turn-on of the synchronous transistor (syncFET) immediately after its body diode recovers.
This phenomenon causes additional switching losses and in the worst cases even a destructive shoot through condition. The effects of both the reverse recovery and the spurious turn-on become more severe with higher voltage and current levels, faster switching transitions, and higher switching frequencies.
The ever more demanding requirements for compact, lightweight and small-sized power converters are pushing their operating frequencies to the values where the sizes of the passive components in the circuit can be significantly reduced. The main negative effect of operating at such high switching frequencies is the increase in switching losses and the resulting lower power conversion efficiency. In order to overcome this drawback, zero-voltage switching (ZVS) approaches are widely applied to all kinds of switching power converters. The utilization of ZVS alleviates the switching losses and significantly decreases the rate of power conversion efficiency deterioration with increasing switching frequency. As a consequence, much higher power densities can be achieved at unaffected power conversion efficiencies.
This paper focuses on switching transitions under zero voltage conditions in a DCM-operated synchronous dc- dc converter. For practical realization of the desirable switching transitions a control strategy is proposed for a conventional nonisolated bidirectional dc-dc converter.
The strategy is based on operation in the DCM and adaptation of the switching frequency. The latter ensures a sufficient magnitude of the reversed current in each switching cycle. In combination with properly selected dead times such control strategy ensures reliable zero- voltage switching transitions of both transistors in buck and boost operating modes within the entire operating range of the converter. In addition, all the spurious turn- on and body diode reverse recovery related issues of the syncFET are avoided. The proposed approach is designed for a conventional bidirectional synchronous
dc-dc converter depicted in Fig. 1. It does not require any auxiliary circuitry or components and can be entirely implemented within a DSC. Due to operation in the DCM, the power inductor’s inductance can be significantly smaller than the one required for the CCM operation in the same operating point. On the other hand, operation with ZVS allows for increasing the switching frequency without a negative effect on the power conversion efficiency. Therefore, a combination of DCM and ZVS offers high potential for reducing the size of the power inductor. Another benefit in terms of power density is a reduction in the size of the heat sink, which is possible due to lower power dissipation on the transistors when operating with ZVS.
II. ZVS IN DISCONTINUOUS CONDUCTION MODE
Switching losses usually represent the highest share in total power loss of a high-voltage low-current hard- switched synchronous dc-dc converter. One way of achieving ZVS in a conventional dc-dc converter that utilizes synchronous rectification is by operating the converter in the DCM. The power circuit of the discussed bidirectional dc-dc converter is shown in Fig.
1. Such a converter can operate in two operating
Fig. 1 : Schematic circuit diagram of a conventional nonisolated bidirectional dc-dc converter.
modes, depending on the direction of the main energy flow. When the average inductor current IL;avg flows in the direction from VA to VB the converter operates in buck mode, in which SH acts as a mainFET and SL as a syncFET. In the other case, when IL;avg flows in the opposite direction, the roles of both transistors in the circuit are exchanged. Fig. 1 also depicts the voltage dependent non-linear parasitic output capacitors CossH and CossL which are of significant importance when studying the switching transitions of power MOSFETs.
The mechanism for achieving zero-voltage turn-on of both transistors in the circuit can be explained on the basis of the equivalent circuits shown in Fig. 2. The explanation, as well as the presented equivalent circuits, is focused on the buck operating mode. However, the same switching states and mechanisms apply for boost mode as well; it is just that the transistors switch roles and the inductor current iL flows in the opposite direction. Each equivalent circuit in Fig. 2 represents
one of eight different switching states inside one switching cycle of a synchronous buck converter operated in DCM. It should be noted that the equivalent circuits (a)-(d) are common to both CCM and DCM operation. After SH is turned off, the main transition period begins that corresponds to the equivalent circuit (b). Assuming that the dead time is
Fig. 2. Equivalent circuit diagram for buck mode
long enough, the inductor current iL completely discharges CossL, while CossH is simultaneously charged to the input voltage VA. As soon as the drain- to-source voltage vdsL across SL falls to zero, the body diode of the syncFET takes over the current, as depicted by circuit (c). In the next phase, the syncFET SL is naturally turned on at zero voltage in both CCM and DCM. The antiparallel body diode of the mainFET SH does not take over any current during this transition and therefore the absence of reverse recovery effect in the mainFETs of the synchronous dc-dc converters.
The idea behind achieving zero-voltage transitions in the DCM is in ensuring the required amount of reversed current IR that can discharge the parasitic output capacitor CossH of the mainFET when operating in buck mode. The principle is evident from the equivalent circuits (e)-(h) in Fig. 2. The turn-off instant of SL is delayed, so that the inductor current iL changes direction, as depicted by equivalent circuit (e). The latter is the first circuit state not present in the CCM or the CCM/DCM boundary operation and is essential for achieving the conditions required for zero-voltage turn- on of the mainFET SH. As the syncFET SL remains in the conductive state, the inductor current iL keeps falling until it reaches the predetermined reversed current IR required for a complete discharge of the parasitic capacitor CossH. After SL is turned off, the main transition period begins. The parasitic output capacitor CossL of the syncFET is charged to the input voltage VA, while the parasitic output capacitor CossH of the mainFET discharges. As soon as the drain-to- source voltage vdsH across SH falls to zero, the antiparallel body diode of this transistor takes over the current, as depicted by equivalent circuit (g).
Subsequently, SH can be turned on at zero voltage. In addition to negligible turn-on losses of SH, the switching states described with equivalent circuits (e)- (h) also avoid passing the current through the antiparallel body diode of SL and thus eliminate the reverse recovery induced losses.
III. ZERO-VOLTAGE TURN-ON OF THE MAINFET
The key for achieving zero-voltage turn-on of the syncFET in both CCM and DCM is in providing a sufficiently long dead time, i.e. time interval between turning off the mainFET and turning on the syncFET.
As the mainFET is turned off at the peak inductor current, which charges its parasitic output capacitance to the voltage VA and discharges the parasitic capacitor across the syncFET in a rather short time, a close to minimum required dead time for preventing simultaneous conduction of both transistors is sufficient for achieving zerovoltage turn-on of the syncFET. In addition, an excessive dead time does not represent a threat of losing ZVS, as there is no mechanism that would cause the reversed process of recharging the parasitic capacitor across the syncFET after it is once discharged. An excessive dead time may only cause a
slight increase in total conduction losses due to a longer conduction period of the mainFET’s body diode.
A. Reversed current considerations
In order to achieve zero-voltage turn-on of the mainFET, the turn-off instant of the syncFET has to be delayed to the point at which the inductor current iL reaches a certain value in the direction that opposes the primary energy flow in a synchronous converter. This value is designated as the reversed current IR. The selection of its value is of crucial importance since an insufficient IR cannot provide enough charge to completely discharge CossH and charge CossL to the voltage VA.
Consequently, a zero-voltage turn-on of the mainFET is not achievable. On the other hand, excessive IR results in an increased inductor current ripple, which leads to higher magnetic core losses, as well as higher turn-off losses of both transistors in the circuit. Although zero- voltage turn-on of the mainFET is maintained in this case, any unnecessary increases in turn-off and magnetic core losses should be avoided. For the aforementioned reasons, a detailed circuit analysis of the main transition period was done.
The inductor current iL(t) can be written as
iL(t)=CossL(vdsL)dvdsL(t)/dt -CossH (vdsH) dvdsH (t)/dt (1)
where vdsL(t) and vdsH(t) are the drain-to-source voltages across transistors SL and SH, respectively, and CossH(vdsL) and CossL(vdsL) voltage-dependent parasitic output capacitors of transistors SL and SH, respectively. The relation between drain-to-source voltages across each transistor can be written as
vdsH(t) = VA � vdsL(t) (2)
where VA is the converter’s input voltage when operating in buck mode. Substituting (2) into (1) and assuming time invariant input voltage VA, which will result in dva/dt = 0, yields
iL(t) = [CossL(vdsL) + CossH(vdsL)] dvdsL(t)/dt (3) where CossH(vdsL) represents the parasitic output capacitance CossH as a function of the complementary transistor’s drain-to- source voltage vdsL.
An expression for the minimum required inductance Lmin for achieving ZVTs at a desirable IR is obtained as,
Lmin =CeqVA(VA - 2VB)/IR2 (4)
IV. EXPERIMENTAL RESULTS
The converter was operating in buck mode with an input voltage of VA = 400 V, while the output voltage was controlled at VB = 200 V.
Table 1. Main components and parameters of the experimental bidirectional DC-DC converter
Symbol description Value
L HSW inductor 660µH
C Capacitor 40µF
F Frequency 115KHZ
VA Voltage 400V
VB Voltage 200V
VA/ VB Duty ratio 0.5
R Resistance 40Ω
Fig 3. ZVS in buck mode of operation
Fig 4. ZVS in boost mode of operation
Fig 5. Closed loop buck operation
Fig 6. Closed loop boost operation
V. CONCLUSION
This paper has proposed a digital control scheme for achieving ZVS throughout the entire operating range of a conventional bidirectional dc-dc converter. The main idea is in ensuring continuous operation of the converter in the DCM with a constant reversed current of sufficient amplitude, which is achieved by altering the switching frequency in accordance with the current load conditions. Such operation eliminates the turn-on losses of both transistors in the circuit and at the same time avoids any reverse recovery and spurious turn-on related issues of the syncFET, regardless of operation in buck or boost mode. The presented experimental results have confirmed the effectiveness and superior performance of the proposed approach in comparison with HSW operation in CCM. Despite operating at higher switching frequencies, the proposed ZVS approach achieved higher power conversion efficiency. Operating in DCM at higher switching frequencies allows for reducing the size of the power inductor and also, to some degree, the
capacitors in the circuit, while a more convenient power loss distribution alleviates the cooling requirements and allows for continuous operation at higher power levels.
Another advantage of the proposed approach is that it does not require any additional components in the circuit of a conventional bidirectional dc-dc converter and can be entirely implemented within a DSC.
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