The system and the transmitter
2.2 Components of the radar
The following description concentrates on big-ship radars, which lie between the large VTS sets and small-craft radars.
2.2.1 Transmission
Each transmitted pulse is a pulselength burst of sinusoid having the very high frequency necessary for efficient propagation close to the sea surface, typically 3 or 9.4GHz (3 or 9.4 x 109Hz). Corresponding wavelength is 10 or 3.2 cm, much shorter than conventional radio practice. The radar is therefore said to operate at microwave frequency or centimetric wavelength. The microwave sinusoid is the carrier (or bearer) of frequency /c, modulated by a train of rectangular unidirectional baseband or video pulses, shaped as Figure 2.9(a), at the prf frequency /m, typically 1 kHz. The microwave magnetron power oscillator is switched on for the duration of each pulse by a modulator device. Modulation superimposes the train on the carrier;
Figure 2.9(b).
Speaking generally, although the energy of a sine wave signal is concentrated at a single (fundamental) frequency, all pulse trains have energy components spread between the fundamental (the prf) and its harmonics. A pulse train, prf = / = I/T, having pulses of any desired shape, can be synthesised by summation of a judiciously chosen d.c. component, plus a Fourier series of sine waves of frequencies / , 2 / , 3 / , . . . , nf, of appropriate amplitudes and phasing. (It is permissible to speak of phasing of these differing frequency components because they are harmonically related.) Where the ratio of the pulse on time, r, to the pulse repetition interval is k = r/T9 the frequency of the nth harmonic, /n, is
f
n=nf=
n- =
n±. (2.2a)
As the radar modulation is a pulse train rather than a sinusoid, /m is a spectrum of prf harmonics centred on /c, Figure 2.10, with two equal and opposite sidebands at frequencies
(Zc+ /m) and ( / c - /m) . (2.2b)
Figure 2.9 Echo spectrum. In frequency domain, lower and upper sideband voltages are mirror images centred on carrier frequency /c
Each sideband contains half the pulse energy so occupied bandwidth is doubled to ±.0.5/x as shown in the transmitted spectrum envelope of Figure 2.9(d). Short pulses, especially those having sharp edges, of necessity occupy a wide spectrum.
Energy density (watts per hertz) in the far skirt regions, although low, may be enough to interfere with users at other frequencies. Imperfections in the magnetron may introduce unwanted further spectral components.
Occupied modulated bandwidth = - . (2.2c) The baseband spectrum of Figure 2.9(c) has bandwidth extending from zero to0.5/r.
Occupied baseband bandwidth = — Hz. (2.2d) Conversion between time (Figures 2.9(a) and (b)) and frequency (Figures 2.9(c) and (d)) domains is possible using mathematical Fourier transforms. The harmonic voltages of a rectangular pulse of height £ R are given by an infinite series comprising a d.c. term JCER9 fundamental and harmonics.
2 i
V = &£R H—/SR Y^ - sin nnk cos nx, (2.2e) n ^ n
n—\
where x = 2nT.
Radar pulse trains have k so low (~0.001 max) that the d.c. term can be neglected.
The amplitude of the nth harmonic, Vn (of frequency fn = nk/x Hz) is found by
Microwave frequencies Video (baseband) frequencies
Frequency (a) Baseband pulses Time domain
(b) Pulses ofRF Time domain
Pow< T densi y
Occupied bandwidth, rectangular pulses (c) Rectangular pulse at baseband
Envelope as (a)
(d) Rectangular modulation on carrier Half energy in each sideband Short pulses occupy a wide spectrum Occupied bandwidth, rectangular pulses
Lower sid ?band Mirror image
Upper iideband Copies baseband
Figure 2.10 Radar block diagram and signal flow. Time and frequency domains.
Typical 9 (and 3) GHz band frequencies indicated. Block diagram (a) represents the usual non-coherent system
Frequency, MHz IF signal sidebands reversed
Image frequency signal (d) Reception of image frequency
Baseband signal after second detector Frequency, MHz
IF signal after mixer and bandpass filter Spectrum truncated to IF bandwidth
Local oscillator offset by IF frequency White noise
(c) Frequency domain - reception of echoes Incoming echo (as Figure 2.9(d)) Narrow filter bandwidth broadens pulse,
causing range uncertainty Time, us Range measurement
T. Demodulator output
Edges affected by filter bandwidth
Repeats after ~1 ms
Bandwidth-limited noise
S. Filter output White noise Noisy echo
R. Receiver input Time delay set by target range Q. Magnetron output (as Figure 2.9(b))
P. Modulator pulse (as Figure 2.9(a)) (b) Time domain
Timing M, IF - Microwave, intermediate frequency
Display Scanner bearing
Free-running local oscillator Receiver
Automatic frequency control
Signal Processing
Digital Band pas;
filter T
Video Demodulator Mixer IF amplifier
Target Scanner
(a) Block diagram Non-coherent system
Self-oscillating magnetron High power pulses
Circulator or duplexer and receiver protection
Low noise amplifier if used Modulator
Low power trigger pulses Pulse
generator Transmitter
setting the cosine term to its maximum value of 1.0:
Vn max = ——— sin(7r/wr). (2.2f)
Figure 2.10(a) is a block diagram of a typical radar, and depicts signals in time and frequency domains. The components are grouped within two or three physi- cal modules. Low-power trigger pulses fire the magnetron via the modulator. The magnetron block has too much power abstracted from it (low 2-factor) to define magnetron frequency exactly. The magnetron output feeds the scanner. Echoes are routed to the receiver and thence to the demodulator, which removes the carrier, leav- ing a baseband or video pulse train similar to that generated by the pulse generator, but with delay proportional to target range and, at a given range, height (voltage) dependent on echo strength. The video train is processed to decide which pulses are likely to represent echoes rather than noise or clutter, then fed to the display for viewing by the operator.
Figure 2.10(b) shows events during a single sweep. The pulse generator delivers a train of trigger pulses at prf near 1000 pps to suit the operator's choice of range scale.
(A slight timing jitter to help suppress interference from other radars is not shown).
The pulses fire the modulator, whose output, P, is a train of powerful (25 kW) pulses at the selected length. The magnetron is 40 per cent efficient and generates 1OkW (40 dBW) bursts of oscillation, Q, centred on the microwave frequency of its resonator in the 3 or 9 GHz bands. The pulsed nature of the transmission causes a fairly broad frequency spectrum to be radiated by the scanner.
2.2.2 Reception
A circulator or duplexer (device routing bidirectional signals) directs the return signal R to the receiver, whose input circuit is preceded by devices to protect its sensitive components from burn-out by the powerful transmitter pulses. The first stage is usually a low noise amplifier working at microwave frequency, which lifts the echo voltage well above unwanted noise injected by later parts of the receiver. Microwave amplifiers are expensive and inconvenient, so the main amplification is done at a lower frequency called the intermediate frequency (IF, 50 MHz).
Figure 2.10(c) shows the frequency relationships within conventional marine radars. The signal R is shifted bodily down to IF, here 50 MHz, at point S by a mixer, sometimes called a first demodulator or first detector, the receiver being a superhet (supersonic heterodyne). In more detail, when the weak microwave signal is super- imposed with a strong sine wave from a continuously running local oscillator, LO, whose frequency is offset from transmitter frequency by the intended IF frequency;
this oscillation beats with the echo. Components are generated at the sum and dif- ference of the two frequencies, the latter being accepted as the IF. A symmetrical arrangement of diodes is used as a balanced mixer, which introduces no LO noise.
Balanced mixers have a noise factor around 8 dB so receivers without LNAs have system noise factors around 9 dB.
2.2.3 Non-coherent system
The LO in the system described is a free-running semiconductor microwave oscillator with output power of a few milliwatts. Its frequency must remain approximately tuned to any drifts in magnetron frequency otherwise the mixer output would drift out of the passband of filters further down the receiver. Tuning is primarily by an automatic frequency control (AFC) circuit which applies a correction voltage proportional to IF frequency error to the LO. The correction is typically derived from the changing phase of the double balanced mixer output. A manual fine tune control and associated tuning indicator are sometimes provided.
Within AFC limits, LO frequency 'does its own thing' - it is not exactly harmon- ically related (is non-coherent) to transmitter frequency. In Figure 2.11 (a) the block diagram is redrawn to emphasise the frequency-determining elements, here shown for the 3 GHz band. While the modulator is firing, the magnetron output (Q) centre frequency is determined by the anode resonator block dimensions, surrounded by a spectrum based on that of the modulator pulse, P. The echo (R) spectrum is more or less identical (target RCS is only slightly frequency-sensitive), although of course echo power is drastically lower.
Extra IF bandwidth has to be retained to cover the residual tuning error, degrad- ing SNR, since noise power is proportional to bandwidth. On long-range scales, where receiver bandwidth is least, residual tuning errors may cause some loss of receiver sensitivity, further spoiling SNR. If the limited range of the AFC is exceeded, it may throw off to a large error, grossly degrading receiver perfor- mance, so the AFC loop must be reset when the magnetron or other component is renewed.
After the LNA, the echo is multiplicatively mixed with the microwave continuous- wave LO oscillation, which preserves the spectrum, shifting it bodily to IF. The echo is amplified in a multistage IF amplifier, containing bandpass filters. Filter centre frequency is the nominal difference between LO and magnetron frequencies and bandwidth has to be wide enough to accept the main components of the pulse spectrum, its occupied bandwidth. The IF output, S, is applied to a diode demodulator where it is rectified to give the baseband video pulse, T, whose spectrum approximates modulator pulse P, with amplitude proportional to echo strength at R. This envelope detection process preserves the envelope of the IF and microwave signals (compare Figure 2.10 S and T).
The non-coherent system just described is wasteful of precious signal because the information resident within the echo phasing is discarded. The following systems improve SNR by preserving echo phase information but are more complicated. Use is currently confined to a few VTS systems.
2.2.4 Coherent-on-receive system
Figure 2.\\{b) depicts one form of coherent-on-receive system, which seeks to retain the cost and efficiency advantages of the magnetron. The transmitter and receiver are basically as the non-coherent system except for the local oscillator,
Figure 2.11 Frequency management strategies. Typical 3 GHz band frequencies.
Most radars are non-coherent
whose frequency is synthesised as follows. A coupler (a pair of parallel wave- guides linked by small slots, or parallel stripline conductors) extracts a sample of the transmitter frequency actually generated, from which the pulse sidebands are then removed. The frequency of this sample is remembered between pulses by either a flywheel oscillator or digitally and is maintained exactly at magnetron frequency.
(a) Non-coherent
Restatement of Figure 2.10(a) with different emphasis AFC keeps LO near one IF away from Tx Automatic frequency control loop
Frequency discrimination
Free-running Diode
demodulator to signal processor Video Spectrum = P Phase information lost
IF amplifier Centre frequency 50MHz
Transmitter Power oscillator Frequency source
Magnetron
Scanner
Power modulator spectrum = P
Duplexer Protection Mixer
Bandwidth wide enough to cover LO frequency error
(b) Coherent-on-receive
Locked to magnetron at each pulse COHO 50.00MHz
Flywheel synchronisation Power oscillator Frequency source Magnetron
Transmitter Coupler
STALO
Power modulator snectrum = P
Mixer
50.00 ± P MHz (IF frequency) Coherent
demodulator to signal
processor In-phase (I) Video
Clock
IF amplifier Centre frequency 50 MHz Bandwidth matched to pulse Quadrature (Q)
Spectrum = P
Phase information preserved within I and Q channels
(c) Fully coherent Frequency source
50.00MHz
Frequency locked to COHO Transmitter Scanner High supply power
watts Transistor _ or TWT Modulator
spectrum = P
TWT or klystron Multiplier
x 59 Clock
to signal processor
In-phase (I)
Quadrature (Q) Spectrum = P
Phase information preserved
Centre frequency 50MHz
Bandwidth matched to pulse Items to right of dashed line differ for active arrays 50.00 ± PMHz (IF frequency) STALO
It is mixed with the output of a coherent oscillator (COHO) which runs at IF centre frequency to give a stable local oscillator (STALO) signal. The figure shows typical frequencies.
After the LNA, the echo is mixed with the STALO to give a pulse spectrum centred on IF centre frequency; this feeds the IF amplifier. Filter bandwidth can be matched to the modulator pulselength without need of additional allowance for LO tuning error. The demodulator is a pair of mixers taking direct and 90° phase- shifted drives from the COHO to give in-phase and quadrature (I and Q) video outputs, preserving both the amplitude and phase information contained in the echo. Successful operation is critically dependent on accurate capture of magnetron frequency during the transmitter pulse, followed by drift-free memory during the relatively long inter- pulse reception phase. Relative to non-coherent operation, integration loss is roughly halved.
2.2.5 Fully coherent system
Figure 2.11(c) depicts a fully coherent system. The COHO is the primary frequency source, multiplication (here by 59 and 60) synthesising the STALO and transmit- ter frequencies, respectively. The continuous-wave transmitter feed is modulated by the pulse at low power (facilitating precise pulse-shaping and control of trans- mitter spectrum), followed by a multistage power amplifier, which replaces the magnetron. The STALO frequency is positively locked one IF frequency away from echoes at all times. The remainder of the receiver follows the coherent-on-receive system. Amplifier-type transmitters are bulky, inefficient and expensive, but give operational flexibility. Although not shown, it is straightforward to stagger trans- mitter frequency from pulse to pulse to decorrelate sea clutter and hence improve detection for a given SNR; or to reduce transmitter power; for short-range opera- tion, for example. The amplifier may contain a travelling wave tube (TWT) feeding a klystron amplifier tube. Both are high voltage thermionic valves. Coherent systems are particularly suited to active array scanners, not in current marine service, see Chapter 16.
2.2.6 Ambiguity; image frequency, prf constraints
It is desirable to maximise the number of sweeps - minimising sweep time - taken into the detection process to maximise the effective SNR and get maximum proba- bility of detection for the chosen false alarm rate. But if sweep time is less than the range delay of the furthest target returning a detectable echo, it becomes uncertain or ambiguous whether a plot is from a close target reflecting the last pulse transmitted, or from a more distant echo of an earlier transmission. Ambiguity can be resolved by severely jittering (staggering) the pri but signal to clutter ratio then tends to suffer.
Operation at relatively low prf is universally preferred, the radar being non-ambiguous out to the maximum instrumented range of the display (on big ships often 96 nmi, 178 km, constraining maximum prf to 840 pps on long-range scales). On short-range scales higher prf can be used. Beside minimising risk of receiving second time around
echoes, relatively low prf is often retained to enable plotting aids to continue to track targets which are too distant for current display, ready for the operator to return to long-range operation; to permit a second display to show the distant scene while the primary display examines the short-range scenario, or to service guard zones.
Low-power radars can utilise high prf without risking significant second-time around echoes from distant targets, except under conditions of anomalous propagation or anaprop, Chapter 5, Section 5.2.5. It is questionable whether the 96 nmi instrumented range frequently provided serves much purpose, only inland mountains being likely to rise above the radar horizon. The operator is sometimes provided with a long/short pulselength switch. Long pulse/narrow bandwidth reduces noise and improves detec- tion of weak targets if clutter is slight; short pulse/wide bandwidth improves range resolution and illuminates less clutter around the target, improving clutter rejection and helping to depict target aspect.
Figure 2.10(d) shows that, depending on detail design, the receiver may be responsive or open to a band of unwanted frequencies lying on the other side of LO frequency from the transmitter. Usually this image frequency band contains nothing synchronised with the transmitter and merely contributes some additional noise, which is included within the overall noise figure.
Swept-frequency racons are sometimes received at image frequency, see Chap- ter 8. Modern receivers include double balanced mixers which reject image frequency.
2.2.7 Typical station configuration
Figure 2.12 shows a typical radar station and its links with other bridge equipment, Figure 2.13 being a particularly futuristic realisation of the bridge components. A pair of transmitter/receivers, if in the same band, may be connected to a single scanner by a combining device called a diplexer (not duplexer), diplex operation being the simul- taneous transmission/reception of two signal channels using a common component such as a scanner. In principle it is possible to combine or fuse the receiver output data streams before feeding a single display. The difficulties exceed the advantages on shipboard, although data fusion is occasionally used in VTS. The bigger ships are mandated by IMO to carry two radars, primarily for reliability. One set has to use the 9 GHz band. IMO encourage the other to be at 3 GHz, giving the operational advantages of each band. Good seamanship usually requires one of a pair of radars or displays to be kept on a long range scale for landfall verification and to give early warning of traffic movement, the other running on a shorter scale for collision avoidance.
Here the installations have their own scanners, but either's receiver output may be switched to the other's display, and it is not always very obvious to the operator which band is active. A very few merchant ships voluntarily duplicate their equipment with typically two each of 3 and 9 GHz radars. Many roll-on roll-off (ro-ro) ferries and some other ships carry additional small, usually 9 GHz, radars low down forward and sometimes astern to assist berthing. Lack of omnidirectional azimuth coverage unsuits them to general navigation.
Figure 2.12 Typical station configuration. Twin-radar installations usually use two scanners which usually remain functionally independent even when sharing a display. The display may contain other facilities, particularly when forming part of an integrated bridge system (IBS)
The remainder of this chapter details how radars illuminate targets. Reception is detailed in Chapter 3.