Feeder - The system and the transmitter - Target Detection by Marine Radar

The system and the transmitter

2.6 Feeder

Combination of the transmitter and receiver input stage with the scanner aloft elimi- nates feeder loss, but may impede servicing access and reduce reliability in the hostile masthead environment. Feeders do not always get proper attention in system design.

At 3 GHz, a coaxial cable is usual, but at 9 and 14 GHz cable is generally too lossy and tubular metal waveguide is preferred.

2.6.1 Waveguide

Figure 2.14 shows an electromagnetic wave propagating energy through a vacuum, air or other dielectric medium. There are mutually perpendicular electric (E) and magnetic (H) fields, the direction of energy flow being mutually perpendicular to both.

When a plane wave such as that of Figure 2.14(a) strikes a perfectly conducting metal sheet placed normal to the direction of propagation,

• the magnetic field component of the wave induces circulating surface currents of finite amplitude in the sheet;

• with zero resistivity, there can be no voltage in the plane of the sheet. So immediately clear of the surface there must be a cancelling electric field in phase opposition to the incident wave.

These conditions make the sheet re-radiate as an antenna and the incident beam is specularly reflected just like light at a mirror. The angles of incidence and reflection are equal. Ray paths can be drawn geometrically, the rays travelling in straight lines when the medium is uniform. (Sometimes, e.g. when a ray grazes an obstacle and diffracts around its surface, behaviour ceases to be properly represented by geometrical optics and can only be described by the much more complex rules of wave theory.)

Figure 2.15 shows a plane wave reflecting from a flat conducting metal sheet.

The angles of incidence and reflection are each 0. Phase fronts are shown, denoting Next Page

transmitter and receiver. Each component introduces some loss. The radar's datasheet power may mean: (a) power at the magnetron output, or (b) the available power at the flange of the transceiver cabinet or at the scanner flange when the scanner is integral with the transmitter. Unless specified otherwise it is prudent to assume (a) - the highest value in the datasheet! - so allowance must be made for internal loss, typically 0.5 dB. Feeder loss is specific to the installation and always excluded from datasheet ratings.

Transmitter power is only one of the parameters affecting performance. Although 25 kW sounds grander than 20 kW, it represents barely 1 dB more (44, 43 dBW).

For equal performance it might be cheaper to choose a set with receiver noise figure 1 dB lower, or to shorten the path of the feeder to reduce its one-way loss by 0.5 dB (see below). Do not be fooled into believing transmitter dBW are somehow more important than dBs elsewhere.

2.6 Feeder

At 3 GHz, a coaxial cable is usual, but at 9 and 14 GHz cable is generally too lossy and tubular metal waveguide is preferred.

2.6.1 Waveguide

When a plane wave such as that of Figure 2.14(a) strikes a perfectly conducting metal sheet placed normal to the direction of propagation,

• the magnetic field component of the wave induces circulating surface currents of finite amplitude in the sheet;

• with zero resistivity, there can be no voltage in the plane of the sheet. So immediately clear of the surface there must be a cancelling electric field in phase opposition to the incident wave.

Figure 2.15 shows a plane wave reflecting from a flat conducting metal sheet.

The angles of incidence and reflection are each 0. Phase fronts are shown, denoting Previous Page

Figure 2.14 Propagation, (a) Shows the electric (E) and magnetic (H) fields of a horizontally polarised ray. Reversing polarity of either field reverses the direction of propagation, (b) is for vertical polarisation. Vector addition of (a) and (b) gives slant polarisation. Vector addition of (a) and (b) with JT/'2 rad phase shift gives circular polarisation, the hand depending on which planar component leads

positions having common phase and one wavelength apart. They are always normal to the direction of propagation. Clearly, once a pattern of incident and reflecting fronts has been launched, a second conducting sheet could be inserted parallel to the first sheet at position YY (spaced distance a) without affecting the pattern in the bounded space. The ray perpendicular to the fronts represents the oblique direction of propagation of the plane wave, assumed to have been launched from some point to the left of the diagram. This wave reflects back and forth as it proceeds towards the right within the bounded space. The velocity in the oblique direction is that of light, c, so the horizontal component of velocity is lower and depends on the obliquity. So once a ray has been launched between two parallel conducting plates, it propagates freely.

Electric field exists between the plates, while magnetic fields link with circulating currents on their inner surfaces.

This geometry requires a whole number of projections of a half-wavelength between the plates, therefore sin 6 = nXo/(2a), where n is an integer. The following waveguide properties arising from this geometry can be proved rigorously from Maxwell's fundamental propagation equations, see, for example, Mehler [2].

The plates can be bounded by another pair, giving a rectangular cross section, making a pipe called a waveguide. The distance between this second pair, b, affects the characteristic impedance (discussed later), and is usually made about a/2, partly

Horizontal H field Observer

Vertical (b) Vertical polarisation

E field Phase fronts

normal to direction of propagation HorizontalE field

Electromagnetic axis

Propagation //field

Vertical (a) Horizontal polarisation

Figure 2.15 Reflection from flat sheet, and propagation in waveguide. Properties can be described using geometrical optics, with total reflection at the narrow walls

to inhibit unwanted transmission modes in the wrong plane. The desired mode of transmission is then called the Hoi mode. If frequency is too high, over-moding may occur, unwanted high-order modes existing as if the guide were compartmented into a honeycomb of parallel smaller tubes. The electric field is everywhere perpendicular to the broad face, varying sinusoidally from zero at the sides to a maximum at the centreline. The magnetic field forms closed loops. Intensities at any given point vary sinusoidally with time.

The pattern of electric and magnetic fields travels along the guide. Its phase velocity is faster than the speed of light:

phase velocity = , (2.4a)c

Vi - UlU)

From the geometry, as signal wavelength rises, the ray (velocity c) becomes more oblique and forward progress is slower. Eventually, when Xo = 2a, 6 reaches it /2 rad, perpendicular to the guide axis, making propagation impossible. Below this cut-off frequency, /c, a given height waveguide will not propagate but reflects the signal back to the source. The velocity of information propagation in the guide is called the group velocity and is the forward component of the velocity vector. It is lower than

Ray normal to phase fronts propagates obliquely at speed of light, c (b) Waveguide

Cross section showing electric field Current loops but no voltage at conducting surface

(a) Ray reflected from plate

Reflecting plate at Y Y would not disturb pattern Phase fronts

normal to axes of rays Positive fronts, solid line Negative fronts, dotted line

Reflected ray

Angle of incidence = angle of reflection Incident

ray

Horizontally polarised Electric field normal to page Flat conducting metal plate

Phase reversal at reflection

Incident and reflected fronts form pattern (see b below)

the velocity of light. From the geometry;

group velocity = c x y l - (X0/2a)2. (2.4b)

Within the guide, wavelength, Ag, is increased relative to free-space wavelength Ao.

It follows from the geometry that:

= CAo⁼ ^o^(2Ac)

group velocity J\ _ (AD/2a)2

For example, at 9.4 GHz (A⁰ = 3.136 cm), waveguide WG16 (WR90, RlOO) having a = 22.86 mm has group velocity = 0.728c = 218.1 m/|xs and Ag = 4.448 cm.

Waveguide, but not coaxial cable, is dispersive; the velocity of propagation varies with frequency. If the pulse has a wide spectrum (short pulselength), dispersion in a lengthy waveguide operating not much above cut-off may spread the echo between a couple of range cells in the data extraction system, reducing detectability. In more normal circumstances dispersion can be ignored. For example, 9300MHz (Ao = 0.03216m) in WG16 has group velocity 213.226 m/|xs, taking 468.94ns to transit a 100 m feeder. Changing to 9500MHz, the velocity rises to 216.922 m/|xs, reducing transit time by 7.94 ns to 460.99 ns, spreading the echo over 1.2 m range equivalent.

Any mismatch at the scanner changes phase by nearly 514rad at the magnetron, causing the 'long line effect' referred to in Section 2.3.2.

The cut-off frequency has free-space wavelength Ao = 2a. Cut-offsets a definite lower limit to the workable frequency. Immediately above cut-off, operation is very frequency dependent. On the other hand, if frequency is too far above cut-off, over- moding occurs. These factors define the waveguide cross section, chosen from one of a range of standard sizes, see Table 2.2. In WR nomenclature, the number is the broad internal //-plane dimension a, expressed in units of 0.01 inch = 0.254 mm.

Waveguide has only about 15 per cent useful bandwidth, which is however sufficient to cover a marine band. Preferred operating frequency ~1.5 x cut-off. Velocity of propagation in coaxial or waveguide feeders is usually about three-quarters the speed of light.

Current flow is confined to the inner surface, the small skin depth (limited by electromagnetic interactions within the metal) precluding any exterior voltage. Waveguide tubes may be rigid rectangular or semi-rigid elliptical in section. The elliptical semi- rigid type has few joints and is less susceptible to water entry. Although too expensive and lossy for long runs, fully flexible and twistable/flexible short lengths are useful to accommodate misalignments and vibration.

Peak power rating is set by arcing voltage. Higher frequency waveguides have smaller cross section, with more loss and lower rated power. In theory, at atmospheric pressure size WG 10 for 3 GHz breaks down at 2.6 MW and WG 16 (9 GHz) at 256 kW, but it is imprudent to operate at more than half breakdown power. Bends generally withstand lower power than straight lengths and need particular consideration on 9 and 14GHz high-power VTS installations. Pressurised dry air or nitrogen is sometimes used to obviate water ingress and increase flashover voltage. Flashover may severely

Table 2.2 Feeder sizes and ohmic loss, copper plain rectangular waveguide, one-way

Band 3GHz 9GHz 9GHz, 14GHz low loss

Waveguide size WGlO, R32, WG16,R100, WG15,R84, WG18,R140, (see note) WR284 WR90 WRl 12 WR62 Outside 76.2x38.2 25.4 x 12.7 31.75x15.875 17.83x9.93

dimensions (mm)

Internal, 72.14 x 34.04 22.86 x 10.16 28.70 x 12.83 15.8 x 7.899 HxE planes (mm)

Loss: plain straight 0.02 0.11 0.09 0.27 guide (dB/m)

Loss: likely in 0.04 0.18 0.15 0.35 service (dB/m)

Note: Waveguide sizes quoted are RCSC, IEC-R 153 and RETMA respectively and are derived from inch units.

damage both transmitter and receiver. In marine radar service, power handling is rarely a problem, unless bad mismatch is introduced by a dent, broken joint or water ingress.

When using coaxial cable, high power necessitates large diameter with substantial bending radius.

2.6.2 Mismatch

Mismatch occurs whenever energy is fed from one component to another of differing impedance. All transmission lines have characteristic impedance, Zo, a vector quan- tity. For coaxial cable Zo ~ ^fLjC where L is the central conductor's inductance per unit length and C is the capacitance per unit length between the central conductor and the metal sheath. Figure 2.16 shows these distributed constants conventionally, as a series of lumped elements. Although less simply defined, waveguides also have characteristic impedance, rising with the b dimension. The device at the far end of the feeder is called the load. The load on a transmitter is the scanner; to echoes it is the receiver. If the load is resistive and equals Zo, it accepts all the energy flowing along the feeder. None is wasted or rejected so the return loss is infinite. This is the desired matched load condition.

The load impedance may differ from Zo. One limiting case is a short-circuit.

At the shorted load terminals current flows but no voltage can exist. Power = VI and no power is accepted as V = 0. The power rejected must go somewhere; it is reflected back toward the source. The load is mismatched. A quarter-wave back along the feeder the incident and reflected voltage components are in phase and add to give a high-voltage node. Slightly further back they are in opposite phase giving a low- voltage antinode. If we measured the voltage along the feeder we should find alternate

Perfect match VSWR Poor match

Figure 2.16 Feeder mismatch. Transmission and return losses plotted against VSWR. Within waveguides, electric fields are in the E plane and magnetic fields in the H plane (mnemonic: easy and hard to bend!)

high- and low-voltage points. A standing wave exists, as when a train of sea waves hits a harbour wall. For our short circuit, the ratio between the node and antinode voltages, called the voltage standing wave ratio (VSWR, sometimes denoted S)9 is infinite for a short or open circuit (the former British practice sensibly used the reciprocal, giving VSWR between 0 and 1). For a matched load VSWR falls to 1.0. VSWR is a measure of match quality. Risk of arcing from the high voltages at mismatch nodes may limit the power handling capacity of feeders.

At spot frequency, a mismatch can be cancelled by another mismatch nearby, usually within the component itself, and this is routinely done during manufacture within scanner/rotating joint and other assemblies. It becomes increasingly hard to maintain match over wide frequency bands or wide physical separation of the original and correcting mismatches. In practice, there are always several residual mismatches, spaced many wavelengths apart. Their phasing becomes extremely frequency depen- dant and there is considerable mismatch uncertainty, preventing accurate prediction of mismatch loss.

As bad VSWR prevents some of the incident power getting to the load, it represents a transmission loss, usually tolerably small unless the mismatch is severe. Often more important is the return loss - the reflection into the receiver when a transmitter feeds a mismatched scanner. Losses are related to VSWR through a voltage reflection

Return loss Left-hand axis

Transmission loss Rieht-hand axis

Mismatch Load

Distributed inductive and capacitative reactances Source

Circuit diagram

E plane H plane a

Hollow metal tube Electric field

Live conductor Low-loss dielectric Outer conductor

Cable Waveguide

Feeder

coefficient, p:

VSWR - 1 .

P = V S W R + 1 n U m e n C a 1' ( 2-5 a )

transmission power loss = —10 log(l — p²) dB, (2.5b) reflection or return power loss = —20 log p dB. (2.5c) Figure 2.16 connects loss and reflection with VSWR and depicts these equations.

2.6.3 Feeder losses

Feeder resistive (ohmic) and mismatch losses affect both transmission and reception paths and can significantly degrade overall performance. Unless there is a pre-amplifier or head amplifier aloft, feeder ohmic loss, L, beside attenuating echoes introduce additional noise, often expressed as a noise temperature = (1 — l/L)T^a Kelvin (Ta is absolute ambient temperature, typically 290 K). With no head amplifier, feeder loss precedes the source of noise in the receiver and the degraded SNR cannot be recouped by subsequent amplification. Ohmic loss can be particularly severe where VTS transceivers are at the foot of tall masts. It is all too easy to throw away expensive transmitter power and low-loss receiver performance in long and lossy feeders.

If a lengthy feed is inevitable, consideration should be given to placement aloft of the main parts of the transceiver, including the head amplifier. Feeder losses are one of the factors affecting choice of scanner height, discussed in Chapter 15, Section 15.2.2.

Losses rise if the waveguide becomes damaged or water seeps in. Kinks sharply increase loss and must be avoided.

Table 2.2 includes a rough guide to practical ohmic insertion losses in undamaged, internally dry, copper waveguide and includes an allowance for internal tarnish and a typical proportion of bends. Losses in semi-rigid and aluminium guide are similar.

The UK Maritime and Coastguard Agency³ offers useful hints on good installation practise for feeders.

The following reduce feeder losses.

• Microwave circuit devices such as scanners having good match.

• Feeder cables of the correct characteristic impedance or using matching trans- former sections.

• Protection from accidental damage.

• Short length. Avoid 'round the houses plumbing'.

• Cable or elliptical waveguide not curved to less than its recommended radius.

• Minimal number of waveguide bends.

• Do not attempt to bend or twist straight waveguide, use a special section.

• Never cross waveguide planes.

Survey of Merchant Shipping Navigational Equipment Installations: Instructions for the Guidance of Surveyors, UK Maritime and Coastguard Agency.

Mismatch causes a standing wave in the feeder, VSWR rising as mismatch worsens. For example, when VSWR = 2, the associated voltage reflection coefficient, p, is 0.33 (Eq. (2.5a)) and transmission loss is 0.51 dB (Eq. (2.5b)).

Two-way, this throws away ~1.0dB (26 per cent) of radar performance. Reflec- tion loss is 9.65 dB, so as much as 11 per cent of the transmitter power reflects back into the receiver - a kilowatt or more, hugely bigger than the echoes it is designed to handle. Protection devices may prevent damage, but the receiver will have a bout of severe indigestion, taking several microseconds to recover full sensitivity after transmission, spoiling performance out to a kilometre or so range.

2.6.4 Ringing

Transmitter power reflected by mismatch introduces unwanted ringing clutter. As the receiver input is never perfectly matched, it re-reflects. The reflection bounces back and forth along the feeder before finally petering out, false echoes or ringing appearing each time. At the usual velocity of propagation near 200 m/|xs, a 50 m feed would deliver reflections to the receiver each 0.5 |xs or 75 m range equivalent. These successive strong reflected pulses can mask true echoes, spoiling the radar's short- range performance. Chapter 11, Section 11.8, details how they interact, depending on their spacing.

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