Radar receiver
3.5 IF amplifier, demodulator and video sections
3.5.1 IF section
The intermediate frequency subsystem or strip contains several amplification stages with gain stabilised by feedback, utilising transistor integrated circuits, interspersed with the bandpass filters. The demodulator or second detector converts the signal to a unidirectional video or baseband pulse, which after further amplification feeds the processing system.
For the typical frequencies shown in Figure 3.6(a) and Chapter 2, Figure 2.10, the IF is a line spectrum at 50 MHz. This bearer represents the microwave carrier.
A sideband component of the modulation /m, say +10 MHz on the 9410 MHz centre frequency, would cause an IF sideband at 60MHz, also offset 10 MHz from the IF bearer, confirming that the spectrum and pulse shape of the microwave signal are preserved, shifted bodily down in frequency by the mixing process. There is a tendency to raise intermediate frequency above 50MHz as better amplifier chips become available, as signal processors become more elaborate and when very short pulses may be transmitted.
Echo strength varies very widely between, say, a small buoy at long range and a large ship at short range. Unless the receiver has enough dynamic range, strong signals overload its later stages, causing saturation and unwanted stretching of the pulse, destroying some of the information in the echo and introducing range error.
Very strong echoes may give the receiver such a bout of indigestion that reception is paralysed for a few microseconds, spoiling reception of nearby targets.
Automatic gain control (AGC) sets gain at long range to maintain maximum tolerable noise and clutter, which optimises detection of weak targets, helping to reduce dynamic range to manageable proportions. At short range, even small targets return strong echoes and full sensitivity is not needed. A manual gain control enables the operator to optimise performance on the target of greatest current interest.
Dynamic range is usually improved by inclusion of a non-linear logarithmic amplifier (Section 3.5.7).
3.5.2 Filter
Unless the target is long enough to cover a significant range bracket, the echo reach- ing the radar receiver reproduces the pulse shape and spectrum of the incident signal, at drastically reduced amplitude. All the timing information, essential for determina- tion of target range, is carried in the sidebands. The receiver must therefore deliver substantially all the sideband energy to the demodulator.
When a noisy signal passes through a wideband amplifier the noise is also ampli- fied - SNR can only be reduced by reduction of bandwidth (Pn oc B, Eq. (3.2a));
magnifying a photograph does not improve its contrast. For efficient echo detec- tion SNR must be optimised, using a frequency selective filter to pass frequencies containing some noise but also most of the signal, hence preserving most of the timing information within it, while blocking those frequencies with noise but little signal.
Two or more successively narrower IF band-pass filter stages are usually included, so bandwidth reduction is gradual, reducing the risk of strong signals causing ringing and ghost echoes behind the displayed echo pulse. Stand-alone filter blocks in the signal path are often aided by frequency-dependent feedback within the amplifiers.
If filter bandwidth is too wide, all the frequency components of the signal are indeed accepted but with much noise, noise being proportional to bandwidth. If bandwidth is too narrow, signal and noise are both attenuated, letting the baby out with the bathwater; the pulse is also broadened, spoiling range accuracy. Matched filters in which the attenuation/frequency characteristic matches the pulse spectrum give best signal to noise ratio but may slope the echo pulse edges too much to determine
the exact instant of arrival, introducing navigationally undesirable range error to displayed plots, degrading ARPA or ATA track prediction and demanding extra AFC accuracy. The amplitude/frequency transfer function is the complex conjugate of the Fourier transform (in frequency domain) of the signal pulse shape.
However, there is little loss of SNR when the filter response shape differs consid- erably from the matched case. Extra IF bandwidth also has to be retained to cover the residual tuning error, degrading SNR. On long-range scales, where receiver band- width is least, residual tuning errors may cause some loss of receiver sensitivity, further spoiling SNR. IF bandwidth is therefore often made rather wider than the matched value, some loss of SNR being accepted in return for preservation of as much signal data as possible to better preserve timing information. Such a filter is termed a matching filter. The minimum necessary bandwidth, Bn, which must be retained to preserve the shape of the echo pulse, length r, to permit range determination is similar to Chapter 2, Eq. (2.2d):
B n - » . (3.4) In practice, somewhat wider receiver bandwidth is often used at the expense of a dB or so poorer SNR
• to preserve signal data and avoid loss of ranging accuracy;
• to facilitate operation of twin displays set to differing range scales;
• to ease local oscillator tuning accuracy.
When pulselength is restricted to permit highest available prf on short range scales, IF bandwidth is occasionally made somewhat less than necessary bandwidth.
In system performance calculations, where the filter shape is in general not known, the receiver passband is usually assumed rectangular, called an abrupt or hard filter. Frequency components away from band centre are then credited with too much gain, introducing a bandshape loss in the same fashion to the beamshape loss of Section 3.3.1 (Figure 3.2). Figure 3.7 shows the similar spectra of rectangular and Gaussian (minimum spectral width) pulses and a hard filter response. The beamshape loss is shaded for the rectangular pulse; the Gaussian pulse loss is similar.
For most transmitted pulse shapes the matched filter bandwidth is quite close to 1/r. For purposes of system performance calculation, rather than entering detailed analysis of pulse shape and filter shape, the following simplifying assumptions are usually made, with little error:
• filter is hard, with zero loss in passband and infinite loss at all other frequencies.
Although not manufacturable, the concept is useful;
• signal pulse is rectangular;
• filter output contains all the signal energy;
• filter output contains noise power equivalent to filter bandwidth;
• filter bandwidth is assumed to be 1/r (some prefer to use 1.2/r) unless actual value is known;
Figure 3.7 Filter responses
• reduced noise pick-up just within the passband does not fully offset additional noise just beyond, so filtration introduces a filter weighting loss to account for the earlier approximations; Figure 3.7. Loss is somewhat dependent on assumed bandwidth as well as shape but is between 1 and 3 dB when bandwidth is near 1 / r . 3.5.3 Linear and square-law demodulators
The second demodulator determines whether the IF amplifier output contains a can- didate echo. Figure 3.8(a) shows a diode current/voltage characteristic curve. An alternating voltage applied to the anode gives positive half-cycles at the cathode, neg- ative half-cycles being blocked, rectifying the input signal to d.c. with a residual a.c.
component. Following radio practice, the process is often called detection. We prefer the alternative term demodulation, reserving detection for the overall determination of whether a signal contains a candidate echo. Figures 3.8(b) and (c) are circuit dia- grams of simple half-wave and full-wave demodulators. The latter uses centre-tapped transformer Tl to collect information from negative as well as positive components of the IF waveform. When the a.c. is an IF pulse burst (d), a unidirectional pulse is produced at (e). The a.c. component is removed by a smoothing capacitor as shown at (f). The capacitor accepts charge on peaks and holds the voltage intermediately.
It acts as a low pass filter, inherently introducing a small time delay. Radar second detectors are generally driven well into the diode linear region, and here video output voltage is proportional to microwave signal voltage, forming a linear demodulator.
Direct microwave rectification with no mixer or IF stage gives a video-frequency pulse direct from a microwave pulse. Although inefficient, this may suffice within a racon for direct demodulation of the transmissions of interrogating radars (Chapter 8, Section 8.2.5). Near the origin, diode current rises proportional to V2, so diode
-0.5/T IF frequency 0.5/T Frequency Rectangular pulse
(heavy line) Gaussian pulse
Rectangular approximation Rectangular and Gaussian spectra nearly identical within ± 0.5/T of centre
Filter weighting loss represents difference between vertically and diagonally hatched areas
Power
Figure 3.8 Diode demodulator. The smoothed output forms the video signal rectifiers are called square-law demodulators. Video voltage is proportional to input signal power, typical sensitivity into a 1 k£2 load is 1 mV per 1 |xW input, efficiency approximating 0.1 per cent. Efficiency is improved somewhat by a small forward current bias (~50 |xA d.c), putting the signal on the most curved part of the V/I characteristic and reducing the microwave characteristic impedance to a few hundred ohms, matching the feed impedance. Receivers having direct microwave detection are sometimes called crystal-video receivers. When biassed, typical 9 GHz minimum detectable signal is —72 dBW for 2 MHz video bandwidth.
Demodulators of coherent systems take the form of multiplicative mixers, the output frequency being the difference between IF and COHO frequencies, Chapter 2, Sections 2.2.4 and 2.2.5, Figures 2.11(6) and (c). A pair of demodulators is used, accepting in-phase and quadrature IF and COHO components (derived via 90° phase shifters) and delivering in-phase (I) and quadrature (Q) video components, their joint amplitude being proportional to echo amplitude, while their relative amplitudes convey the signal phase information.
3.5.4 Factors affecting detection
Despite swept gain, the receiver input voltage/output voltage relationship or transfer characteristic must support events when the noise and clutter are at their maximum;
(e) Demodulated echo Vy (light line, Cl not fitted) (f) Smoothed video (heavy line) Cl and Rl decay exponentially Cl makes Vw follow signal positive peaks after first few cycles
(d) Echo pulse Duration 0.05-1.0 us
End Start
(a) Diode
At low voltage
l~V2~P S ign a l s o u r c e
Smoothing capacitor Cl
(b) Half-wave rectifier (c) Full-wave rectifier Amplitude Vs
V, volts
Load Rl Diode Dl Vs Anode Cathode At high voltage I ~V
that is the receiver as a whole must have sufficient dynamic range to support strong as well as weak signals without clipping off (limiting) which destroys the information in large events. As well as provoking reflections from feeder mismatch (Chapter 2, Section 2.6.2), excessively strong echoes or clutter can also drive the receiver into paralysis or induce damped oscillation, dumping or repeating nearby targets. Severity of these overload effects is a matter of detail design and difficult to quantify.
3.5.5 Detection cells
The signal processing system divides the surveillance area into a matrix of detection cells, each covering an area of roughly pulselength x (range x azimuth beamwidth).
Big cells cover more surface area so pick up more sea clutter. They also cover more atmospheric volume, getting more precipitation clutter. Thermal noise is indirectly increased by small cell length, for receiver bandwidth has to be wider for short pulses.
Small cells, matched by narrow pulselength and large scanner aperture (narrow beamwidth) are desirable to:
• give crisp display of weak targets, somewhat improving the operator's perception;
• improve signal to clutter ratio unless the target is big enough to overflow the cell;
• resolve adjacent targets, which otherwise merge; often a main reason for giving VTS stations such large scanners;
• increase position accuracy, necessary for accurate ARPA/ATA track prediction;
• but narrow cells have fewer hits available for integration, especially in the case of fast crossing targets whose angular velocity changes azimuth bearing per scan by more than a cell width.
3.5.6 Effect of range scale selection
At long range in benign clutter, the receiver can be operated at high gain to receive weak echoes from distant targets. Bandwidth is minimised to suppress noise and raise SNR, necessitating maximum available transmitter pulse length to satisfy the necessary bandwidth criterion, some range accuracy being sacrificed. PRF is reduced (a) to give unambiguous operation with only one pulse in flight at a time, (b) to avoid transmitter on/off ratio or duty cycle overload.
Radars have 8-10 range scales, in 2 : 1 ratio. At short range, echoes are stronger and more noise is tolerable, so bandwidth is raised and pulselength reduced to improve range resolution. The reduced SNR reduces the likelihood of reception of very distant echoes, easing the range ambiguity problem and enabling transmitter prf to be raised, partly restoring the duty. The additional pulses per packet partially restore sensitivity by integration, somewhat offsetting the increased noise. The short pulses also illumi- nate less clutter. PRF is not in fact always increased to the fullest possible extent as it is often desirable to retain a long-range scale on the secondary viewing display for early warning of new targets during a coastal passage, restricting prf to avoid ambiguity.
Also the ARPA or ATA must continue surveillance of distant undisplayed targets to maintain their tracks. In severe clutter considerably exceeding noise, it is desirable to minimise range cell size by retaining short pulse/wide bandwidth operation.
Table 3.1 Range scale comparison Range scale setting Long Short Pulselength Long Short Cell size Large Small Range discrimination Poor Good PRF Low High Pulses/scan Low High Receiver bandwidth Low High Receiver noise Low High Signal to noise ratio High Low Signal to clutter ratio Low High
Individual radars differ in their matching of pulselength/bandwidth to range scale, and may include intermediate pulse length/bandwidth steps. Table 3.1 summarises the broad effects on detection performance of the chosen range scale.
3.5.7 Video amplifier
The demodulator restores the signal to baseband. The output instantaneous video voltage is proportional to the IF input voltage. The spectrum of any target is con- verted back to that of the modulator, so necessary bandwidth reverts from the IFs 1/r to 0.5/r. Response need not extend down to d.c. but can roll off at ~50kHz, simplifying detail design and rejecting semiconductor low-frequency flicker noise.
Phase information is only preserved in coherent systems, for which twin I and Q video channels are required. Video amplifiers are usually contained on a couple of semiconductor chips.
A conventional linear amplifier has output voltage proportional to the input volt- age. Negative feedback is employed to improve linearity and stabilise gain. Supposing the dynamic range has to be 100 dB and the maximum output the amplifier can give before overload (i.e. within its dynamic range) is 1V, then a weak echo, 100 dB down, has amplitude only 10 |xV. This voltage range of 105 :1 demands a 17-bit analog to digital convenor (217 = 131072 :1 = 102.3 dB) followed by 17-bit (plus parity bits) processing in the signal processor, which is profligate in computing power.
Usually a logarithmic amplifier or log amp is included. Here output voltage is proportional to a x (logarithm of input voltage), so 10OdB (1010) range causes only 10a : 1 output voltage variation, which can be handled by a smaller processor;
when a = 1,10OdB input change (equivalent to 105 : 1 voltage variation) gives log 105 = 50 output change, which is less than 26 or 64, enabling 6-bit plus parity processing. Logarithmic amplifiers sum the outputs of several successive limiting stages; Figure 3.9(a). They can handle extremely wide dynamic range, but their chips have to be precisely made to avoid error, placing the radar designer at the mercy of the
Figure 3.9 Receiver ancillary circuits showing waveforms, (a) Logarithmic ampli- fier. As the signal voltage increases, the last stage saturates, followed by the others in turn, (b) Fast time constant. Breaks up solid blocks of clutter by extracting changes in signal + clutter level, (c) Pulse length discriminator. Breaks up solid clutter by extracting signals matching transmitter pulselength. Not applicable to large targets
chip supplier. Especially when used with a differentiator, they help to reveal echoes within clutter. If used in a constant false alarm rate (CFAR) system which maximises probability of detection, the first limiter stage must always be saturated by noise. The gain control may actually control the clipping level rather than amplifier gain. The human ear has a logarithmic scale, enabling us to handle an extreme loudness range.
When several displays are driven from the output of the main display's logarithmic amplifier, each may be given its independent gain, swept gain and differentiator controls, which may be set to suit each operator's requirements.
3.5.8 Fast time constant, differentiator
In areas of rough water or precipitation producing weather clutter (sea and particu- larly precipitation clutter), solid clutter may return a substantial and near-equal signal
Video output only when echo width matches delay (c) Pulse length discriminator
Inverter
Coincidence Differentiator
as(b)
Delay
= Tx pulsewidth Video
Fast time constant (FTC) or differentiator
RC ~ pulselength (b) Log FTC
Log amp as (a)
IF input Video Differentiated video
Main logarithmic video output Delay lines
compensate amplifier delays (a) Logarithmic amplifier
and demodulator Limiting amplifier Diode
detector
Summation Limited IF output for AGC etc IF input
Figure 3.10 Differentiator in precipitation clutter Time domain for a single sweep.
Differentiation partially suppresses blocks of clutter to reveal the edges of echo pulses
to all detection cells in the area, masking any echoes they may contain, as shown in Chapter 2, Section 2.1.6, Figure 2.6 rain areas. The echoes can be enhanced by look- ing not at signal amplitude, but its rate of change with time during each sweep. Rate of change is mathematical differentiation and in analog systems is achieved by passing the signal through a fast time constant (FTC) circuit, such as a series capacitor-resistor high-pass network. Alternatively, an analog resistor-inductor combination or a digi- tised functional equivalent may be used. Figure 3.9(b) shows a simple FTC circuit following a log amp. Figure 3.9(c) shows a simple pulselength discriminator, which can enhance detection of small area targets, but which is rarely applicable in marine practice.
Figure 3.10 depicts detection of six targets having differing ranges and echo strengths (a) represents the echoes, E1-E6. Part of the range bracket lies in rather heavy precipitation (b) giving solid clutter. The narrow-band IF signal at (c) shows the echoes, the envelope being rounded by the narrow bandwidth. E3-E6 are buried in clutter; each echo being equivalent to echo S in Chapter 2, Section 2.2.1, Figure 2.10(b). After the second detector (d), similar to T in Figure 2.10(b), the base- band signal's detection threshold has to be set high to reduce probability of false alarm (PFA) to an acceptable level. Waveform (e) shows echo E4 is lost and there are false alarms at Fl and F2. The high threshold completely prevents detection of weak echo El and moderate echo E2 is only just seen, although both lie outside the clutter area.
Time (E = echo, F = false alarm)
(g) Di Terentiated det sctions
(f) Di Terentiated T mecom tant 0.5 El pul jewidth
B ock of clutter
(e) Declared (etections (d) At baseband Threshold (c) Narrow-bend IF (b) Precipitation (a) Ec ioes
Waveform (f) shows baseband signal (d) after differentiation. Peak echo ampli- tude has fallen, but clutter has fallen further permitting a lower threshold (g). All echoes except E4 are now detectable, with a couple of false alarms. Strong echo E6 occupies considerable time, equivalent to radial length. It might represent an islet or a racon response. When the radar is operated conventionally, all parts of the echo are detected and displayed as an axial trace. Engaging the differentiator suppresses nearly everything but the leading edge, perhaps with a faint suspicion of the remainder from enhanced noise, E6'. Display quality is reduced. The extended nature of the echo is no longer apparent, neither can the shape of its paint reveal the aspect of a large target. At the end of the echo, voltage overshoot (not shown) in the baseband amplifier circuit may cause a faint blip which might be mistaken for another target.
The figure is drawn for differentiator timeconstant half a pulselength to emphasise the principle. In practice the timeconstant is usually about one pulselength. All radars provide a differentiator or FTC on/off control, usually with control of timeconstant, giving the operator more scope to balance detectability against display quality. As differentiation degrades echo plots, it should be disengaged in clear conditions. Its value lies in its ability to break up solid clutter. Sea clutter is less solid than that from precipitation - the structure of individual sea-waves is often visible as striations on the display - so differentiation is less effective against sea than precipitation clutter.