2. Build the circuit, and apply the worst-case conditions (lowest battery volt
age and highest load current at the desired output voltage).
3. Select an inductor value until the desired output voltage is reached, using the equations for Lx from Sections 5.3.2 and 5.3.3 as a guide. For step- down applications, select an inductor that will produce an output voltage slightly less than desired (to allow for manufacturing tolerances). Remem
ber that the actual output voltage is set by the R,:R2 ratio (see Figs. 5-12 through 5-15).
4. Test the circuit for load-line regulation, efficiency, and ripple as described
5.3.8 Feedback Compensation
When large values (50 kQ and above) are used for the voltage-setting resis
tors, Rj and R2 of Fig. 5-12, stray capacitance at the feedback input (pin 8) can add a lag to the feedback response. This might destabilize the regulator, increase low- frequency ripple, and lower efficiency. The problem can usually be avoided by min
imizing the stray capacitance at pin 8 and by adding a lead-compensation capacitor of 100 pF to 10 nF. In inverting applications, the capacitor connects between -VOUT
and pin 8. For step-down applications, connect the capacitor between ground and pin 8. (Most applications do not require this feedback-compensation capacitor.)
5.3.9 Power-Transistor Interfaces
The most important consideration in selecting an external power transistor is the saturation voltage at IMAX. This is the voltage across the transistor when switched full-on by the regulator IC. A typical saturation voltage is 0.5 V or less.
The lowest saturation voltage produces the best efficiency. Also, transistors with a high beta (high gain) require less base drive and thus improve efficiency (because they consume less drive power).
When troubleshooting external power or switch transistor circuits, make cer
tain that the transistor is being driven by a clean, sharp-edged waveform. Monitor the waveforms with a scope. To check the waveform output from the regulator IC, disconnect the inductor, and tie the feedback pin to some fixed high point (say the battery input) through a 10-kQ resistor. This will apply a fixed unbalance to the comparator (Cj in Fig. 5-12) and cause the regulator to produce output pulses at the maximum duty cycle without drawing excessive inductor currents.
Check the on-time and off-time of the drive pulses. Look for slow rise times that might cause the power transistor to enter the linear-operating region, rather than the desired switching (full-on, full-off state). There should be no problems with the switching drive pulses in the following circuits, if the recommended part numbers are used. However, it may be necessary to adjust resistor values to get a particular power level or input-output voltage. In that case, it may be necessary to select differ
ent values for Cx and Lx using the equations in Sections 5.3.2 and 5.3.3.
5.3.ΊΟ inverting Medium-Power Interface
Figure 5-17 shows a 4391 regulator driving an external power transistor to provide -24 V (at powers from 250 mW to 1 W) with an input of +5 V. Supply volt
age is applied to the regulator through R3. When the internal switch transistor is turned on, current through R4 is also drawn through R3. This creates a voltage drop from base to emitter of the external switch transistor, turning on the transistor at the switching frequency determined by capacitor Cx at pin 3 (about 27 kHz).
Voltage pulses on the supply lead (pin 6) do not affect circuit operation be
cause the internal reference and bias circuits have good supply-rejection capabili
ties. A power Schottky diode (Motorola MBR030) is used for higher efficiency.
o.VF±:
Figure 5-17. Inverting medium-power interface (Raytheon Linear Integrated Circuits, 1989, p. 9-60)
5.3.11 Inverting High-Power Interface
Figure 5-18 shows a 4391 regulator driving external power transistors to pro
vide power outputs from 500 mW to 5 W. The output voltage is set by the RpR2
ratio times 1.25, as shown in Fig. 5-12.
io.VF
R2
VFB VR VS LX
4391
Cx Gnd
i 1 l l
Figure 5-18· Inverting high-power interface (Raytheon Linear Integrated Circuits, 1989, p. 9-61)
This circuit uses an extra external transistor Q2 to provide well-controlled drive current (in the correct phase) to the power switch transistor Qr The value of R3 sets the drive current to the switch by making the Q2 act as a current source. R4
and R5 must be selected such that the RC time constant of R4 and the base capaci
tance of Q2 do not slow the response time (and affect duty cycle), but not so low in value that excess power is consumed and efficiency suffers. The values chosen for R4/R5 should be proportional to the supply voltage (values shown are for +5 V).
5.3.12 Step-Down Interface
Figures 5-19 and 5-20 show medium- and high-power interfaces, modified to perform step-down operation. The design equations and suggestions for the circuits of Figs. 5-17 and 5-18 also apply to these circuits.
5.3.13 Voltage-Dependent Oscillator
Figure 5-21 shows the 4390 connected in a circuit where switch frequency de
pends on input (battery) voltage. This configuration can be applied to the circuits in this section. As discussed in Section 5.2.10, the trade-off between load-current capa
bility and output ripple can be improved with the circuit of Fig. 5-21. This circuit uses the internal low-battery detector to sense a low-battery voltage condition and decreases the oscillator frequency after a programmed threshold is reached. The threshold is programmed in the same way as the normal low-battery detector, where
rT H = V, REF